WO2014018861A1 - Programmable rf notch filter for envelope tracking - Google Patents
Programmable rf notch filter for envelope tracking Download PDFInfo
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- WO2014018861A1 WO2014018861A1 PCT/US2013/052277 US2013052277W WO2014018861A1 WO 2014018861 A1 WO2014018861 A1 WO 2014018861A1 US 2013052277 W US2013052277 W US 2013052277W WO 2014018861 A1 WO2014018861 A1 WO 2014018861A1
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- frequency
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- notch
- signal
- power supply
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
- H03H21/00—Adaptive networks
- H03H21/0012—Digital adaptive filters
- H03H21/002—Filters with a particular frequency response
- H03H21/0021—Notch filters
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
- H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
- H03F1/0211—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the supply voltage or current
- H03F1/0216—Continuous control
- H03F1/0222—Continuous control by using a signal derived from the input signal
- H03F1/0227—Continuous control by using a signal derived from the input signal using supply converters
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
- H03F3/189—High frequency amplifiers, e.g. radio frequency amplifiers
- H03F3/19—High frequency amplifiers, e.g. radio frequency amplifiers with semiconductor devices only
- H03F3/195—High frequency amplifiers, e.g. radio frequency amplifiers with semiconductor devices only in integrated circuits
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
- H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
- H03F3/24—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages
- H03F3/245—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages with semiconductor devices only
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
- H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
- H03H7/01—Frequency selective two-port networks
- H03H7/0107—Non-linear filters
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
- H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
- H03H7/01—Frequency selective two-port networks
- H03H7/0153—Electrical filters; Controlling thereof
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F2200/00—Indexing scheme relating to amplifiers
- H03F2200/432—Two or more amplifiers of different type are coupled in parallel at the input or output, e.g. a class D and a linear amplifier, a class B and a class A amplifier
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F2200/00—Indexing scheme relating to amplifiers
- H03F2200/555—A voltage generating circuit being realised for biasing different circuit elements
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
- H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
- H03H7/01—Frequency selective two-port networks
- H03H2007/013—Notch or bandstop filters
Definitions
- the present disclosure relates to direct current (DC)-DC converters and circuits that use DC-DC converters.
- DC-DC converters often include switching power supplies, which may be based on switching at least one end of an energy storage element, such as an inductor, between a source of DC voltage and a ground.
- an output voltage from a DC-DC converter may have a ripple voltage resulting from the switching associated with the energy storage element.
- the ripple voltage is undesirable and is minimized as much as sizes and costs permit.
- a parallel amplifier, a switching supply, and a radio frequency (RF) notch filter are disclosed.
- the parallel amplifier has a parallel amplifier output, such that the switching supply is coupled to the parallel amplifier output.
- the RF notch filter is coupled between the parallel amplifier output and a ground.
- the RF notch filter has a selectable notch frequency, which is based on an RF duplex frequency.
- the parallel amplifier and the switching supply combine to provide a first envelope power supply signal to an RF power amplifier (PA), such that the first envelope power supply signal at least partially envelope tracks an RF transmit signal, which is provided by the RF PA.
- PA RF power amplifier
- the RF notch filter reduces noise introduced by the first envelope power supply signal into RF receive circuitry. As such, the selectable notch frequency is selected to target specific receive frequencies.
- Figure 1 shows a direct current (DC)-DC converter according to one embodiment of the present disclosure.
- Figure 2 shows the DC-DC converter according to an alternate embodiment of the DC-DC converter.
- Figure 3 is a graph illustrating a frequency response of a radio frequency (RF) notch filter illustrated in Figure 1 according to an additional embodiment of the DC-DC converter.
- RF radio frequency
- Figure 4 shows the DC-DC converter according to another
- Figure 5 shows the DC-DC converter according to a further embodiment
- FIG. 6 shows a radio frequency (RF) communications system according to one embodiment of the present disclosure.
- Figure 7 shows the RF communications system according to an alternate embodiment of the RF communications system.
- Figure 8 shows the RF communications system according to an additional embodiment of the RF communications system.
- Figure 9 shows the RF communications system according to another embodiment of the RF communications system.
- Figure 10 shows the DC-DC converter according to one embodiment of the DC-DC converter.
- Figure 1 1 shows the DC-DC converter according to an alternate embodiment of the DC-DC converter.
- Figure 12 shows the RF communications system according to one embodiment of the RF communications system.
- Figure 13 shows details of the DC-DC converter illustrated in Figure 12 according to one embodiment of the DC-DC converter.
- Figure 14 shows details of an RF notch filter used in the DC-DC converter illustrated in Figure 13 according to one embodiment of the RF notch filter.
- Figure 15 shows the details of the RF notch filter used in the DC-DC converter illustrated in Figure 13 according to an alternate embodiment of the RF notch filter.
- Figure 1 6 shows the details of the RF notch filter used in the DC-DC converter illustrated in Figure 13 according to an additional embodiment of the RF notch filter.
- Figure 17 is a graph illustrating frequency behavior of the RF communications system illustrated in Figure 12 according to one embodiment of the RF communications system.
- FIG. 1 shows a direct current (DC)-DC converter 10 according to one embodiment of the present disclosure.
- the DC-DC converter 10 includes a switching supply 12, a parallel amplifier 14, and a radio frequency (RF) notch filter 18.
- the switching supply 12 includes switching circuitry 1 6 and a first inductive element L1 .
- the parallel amplifier 14 has a feedback input FBI and a parallel amplifier output PAO.
- the switching supply 12 is coupled to the parallel amplifier output PAO.
- the switching circuitry 1 6 has a switching circuitry output SCO.
- the first inductive element L1 is coupled between the switching circuitry output SCO and the feedback input FBI.
- the RF notch filter 18 is coupled between the parallel amplifier output PAO and a ground.
- the parallel amplifier output PAO is directly coupled to the feedback input FBI, as shown.
- the parallel amplifier 14 partially provides a first power supply output signal PS1 via the parallel amplifier output PAO based on a voltage setpoint.
- the switching supply 12 partially provides the first power supply output signal PS1 via the first inductive element L1 .
- the switching supply 12 may provide power more efficiently than the parallel amplifier 14.
- the parallel amplifier 14 may provide a voltage of the first power supply output signal PS1 more accurately than the switching supply 12.
- the parallel amplifier 14 regulates the voltage, called a first voltage V1 , of the first power supply output signal PS1 based on the voltage setpoint of the first power supply output signal PS1 .
- the switching supply 12 regulates the first power supply output signal PS1 to minimize an output current, called a parallel amplifier output current IP, from the parallel amplifier 14 to maximize efficiency.
- the parallel amplifier 14 behaves like a voltage source and the switching supply 12 behaves like a current source.
- the switching circuitry 1 6 provides a switching output voltage VS and an inductor current IL to the first inductive element L1 via the switching circuitry output SCO.
- the DC-DC converter 10 receives a DC source signal VDC, such that the parallel amplifier 14 partially provides the first power supply output signal PS1 using the DC source signal VDC and the switching supply 12 partially provides the first power supply output signal PS1 using the DC source signal VDC.
- FIG. 2 shows the DC-DC converter 10 according to an alternate embodiment of the DC-DC converter 10.
- the DC-DC converter 10 illustrated in Figure 2 is similar to the DC-DC converter 10 illustrated in Figure 1 , except the DC-DC converter 10 illustrated in Figure 2 further includes power supply control circuitry 20 and the switching supply 12 further includes a filter capacitive element CF.
- the filter capacitive element CF is coupled between the parallel amplifier output PAO and the ground. As such, the filter capacitive element CF may significantly reduce unwanted noise, ripple, or both from the first power supply output signal PS1 .
- the power supply control circuitry 20 receives the DC source signal VDC and is coupled to the parallel amplifier 14 and the switching circuitry 1 6.
- FIG. 3 is a graph illustrating a frequency response 22 of the RF notch filter 18 ( Figure 1 ) illustrated in Figure 1 according to an additional embodiment of the DC-DC converter 10.
- the RF notch filter 18 ( Figure 1 ) has the frequency response 22 with an RF notch 24 at an RF notch frequency RNF. Therefore, the RF notch filter 18 ( Figure 1 ) filters the first power supply output signal PS1 ( Figure 1 ) based on the frequency response 22. As such, the RF notch filter 18 ( Figure 1 ) may significantly reduce unwanted noise, ripple, or both from the first power supply output signal PS1 ( Figure 1 ) at the RF notch frequency RNF.
- the RF notch filter 18 ( Figure 1 ) is a programmable RF notch filter, such that the RF notch frequency RNF is a selectable notch frequency.
- an alternate programmable RF notch filter such that the RF notch frequency RNF is a selectable notch frequency.
- the RF notch filter 18 ( Figure 1 ) is a fixed RF notch filter, such that the RF notch frequency RNF is not a selectable notch frequency.
- the RF notch frequency RNF is equal to about 10 megahertz. In a second embodiment of the frequency response 22, the RF notch frequency RNF is equal to about 20 megahertz. In a third embodiment of the frequency response 22, the RF notch frequency RNF is equal to about 30 megahertz. In a fourth embodiment of the frequency response 22, the RF notch frequency RNF is equal to about 40 megahertz. In a fifth embodiment of the frequency response 22, the RF notch frequency RNF is equal to about 50 megahertz.
- FIG. 4 shows the DC-DC converter 10 according to another embodiment of the DC-DC converter 10.
- the DC-DC converter 10 illustrated in Figure 4 is similar to the DC-DC converter 10 illustrated in Figure 2, except the DC-DC converter 10 illustrated in Figure 4 further includes an offset capacitive element CO coupled between the parallel amplifier output PAO and the feedback input FBI.
- the switching supply 12 is coupled to the parallel amplifier output PAO via the offset capacitive element CO
- the RF notch filter 18 is coupled to the parallel amplifier output PAO via the offset capacitive element CO.
- the RF notch filter 18 includes a notch filter capacitive element CT and a notch filter inductive element LT coupled in series.
- the notch filter capacitive element CT and the notch filter inductive element LT form a resonant circuit having a resonant frequency.
- the RF notch frequency RNF ( Figure 3) is based on the resonant frequency.
- a shape of the frequency response 22 ( Figure 3) near the RF notch frequency RNF ( Figure 3) may be based on a Q factor of the resonant circuit.
- the parallel amplifier 14 partially provides the first power supply output signal PS1 via the parallel amplifier output PAO and the offset capacitive element CO based on the voltage setpoint.
- the offset capacitive element CO allows the first voltage V1 to be higher than a voltage at the parallel amplifier output PAO.
- the parallel amplifier 14 may properly regulate the first voltage V1 even if the first voltage V1 is greater than a maximum output voltage from the parallel amplifier 14 at the parallel amplifier output PAO.
- the filter capacitive element CF is coupled between the parallel amplifier output PAO and the ground through the offset capacitive element CO.
- the offset capacitive element CO is omitted.
- FIG. 5 shows the DC-DC converter 10 according to a further embodiment of the DC-DC converter 10.
- the DC-DC converter 10 illustrated in Figure 5 is similar to the DC-DC converter 10 illustrated in Figure 4, except in the DC-DC converter 10 illustrated in Figure 5, the RF notch filter 18 includes the notch filter capacitive element CT, the notch filter inductive element LT, and a notch filter resistive element RT coupled in series.
- the notch filter capacitive element CT, the notch filter inductive element LT, and the notch filter resistive element RT form a resonant circuit having a resonant frequency.
- the RF notch frequency RNF ( Figure 3) is based on the resonant frequency.
- a shape of the frequency response 22 ( Figure 3) near the RF notch frequency RNF ( Figure 3) may be based on a Q factor of the resonant circuit.
- FIG. 6 shows a radio frequency (RF) communications system 26 according to one embodiment of the present disclosure.
- the RF radio frequency
- the communications system 26 includes RF transmitter circuitry 28, RF system control circuitry 30, RF front-end circuitry 32, an RF antenna 34, and a DC power source 36.
- the RF transmitter circuitry 28 includes transmitter control circuitry 38, an RF power amplifier (PA) 40, the DC-DC converter 10, and PA bias circuitry 42.
- the DC-DC converter 10 functions as an envelope tracking power supply.
- the DC power source 36 is external to the RF communications system 26.
- the RF front- end circuitry 32 receives via the RF antenna 34, processes, and forwards an RF receive signal RFR to the RF system control circuitry 30.
- the RF receive signal RFR has an RF receive frequency.
- the RF notch frequency RNF ( Figure 3) is about equal to the RF receive frequency, which may reduce noise, ripple, or both in the receive path from the transmit path or other noise sources.
- the RF system control circuitry 30 provides a power supply control signal VRMP and a
- the transmitter configuration signal PACS to the transmitter control circuitry 38.
- the RF system control circuitry 30 provides an RF input signal RFI to the RF PA 40.
- the DC power source 36 provides a DC source signal VDC to the DC-DC converter 10.
- the DC power source 36 is a battery.
- the power supply control signal VRMP is an envelope power supply control signal.
- the DC power source 36 provides the DC source signal VDC to the parallel amplifier 14 ( Figure 1 ) and to the switching supply 12 ( Figure 1 ).
- the transmitter control circuitry 38 is coupled to the DC-DC converter 10 and to the PA bias circuitry 42.
- the DC-DC converter 10 provides the first power supply output signal PS1 to the RF PA 40 based on the power supply control signal VRMP.
- the DC-DC converter 10 is an envelope tracking power supply and the first power supply output signal PS1 is a first envelope power supply signal EPS.
- the DC source signal VDC provides power to the DC-DC converter 10.
- the first power supply output signal PS1 which is the first envelope power supply signal EPS, is based on the DC source signal VDC.
- the power supply control signal VRMP is representative of a voltage setpoint of the first envelope power supply signal EPS.
- the RF PA 40 receives and amplifies the RF input signal RFI to provide an RF transmit signal RFT using the first envelope power supply signal EPS.
- the first envelope power supply signal EPS provides power for amplification to the RF PA 40.
- the first envelope power supply signal EPS is amplitude modulated to at least partially provide envelope tracking.
- the RF PA 40 operates with approximately constant gain, called isogain, and with gain compression.
- the gain compression is greater than about one decibel.
- the gain compression is greater than about two decibels.
- the gain compression is equal to about two decibels.
- the gain compression is equal to about three decibels.
- the gain compression is equal to about four decibels.
- a bandwidth of the first envelope power supply signal EPS is greater than or equal to about 10 megahertz. In a second embodiment of the first envelope power supply signal EPS, a bandwidth of the first envelope power supply signal EPS is less than or equal to about 10 megahertz. In a third embodiment of the first envelope power supply signal EPS, a bandwidth of the first envelope power supply signal EPS is greater than or equal to about 20 megahertz. In a fourth embodiment of the first envelope power supply signal EPS, a bandwidth of the first envelope power supply signal EPS is less than or equal to about 20 megahertz.
- the RF front-end circuitry 32 receives, processes, and transmits the RF transmit signal RFT via the RF antenna 34.
- the transmitter control circuitry 38 configures the RF transmitter circuitry 28 based on the transmitter configuration signal PACS.
- the RF communications system 26 operates in a full duplex environment, such that the RF transmit signal RFT and the RF receive signal RFR may be active simultaneously.
- the RF transmit signal RFT has an RF transmit frequency and the RF receive signal RFR has the RF receive frequency.
- a difference between the RF transmit frequency and the RF receive frequency is about equal to an RF duplex frequency.
- the RF notch frequency RNF ( Figure 3) is about equal to the RF duplex frequency, which may reduce noise in the receive path from the transmit path.
- the RF notch filter 18 is a programmable RF notch filter, such that the RF notch frequency RNF is a selectable notch frequency.
- the selectable notch frequency is based on the RF duplex frequency.
- the selectable notch frequency is about equal to the RF duplex frequency.
- the transmitter control circuitry 38 selects the selectable notch frequency based on the RF duplex frequency.
- the RF system control circuitry 30 provides notch frequency information to the
- the transmitter control circuitry 38 via the transmitter configuration signal PACS.
- the notch frequency information is based on the RF duplex frequency. Then, the transmitter control circuitry 38 selects the selectable notch frequency based on the notch frequency information. As such, selection of the selectable notch frequency is based on the notch frequency information.
- the RF duplex frequency is greater than or equal to about 10 megahertz. In a second embodiment of the RF duplex frequency, the RF duplex frequency is greater than or equal to about 20 megahertz. In a third embodiment of the RF duplex frequency, the RF duplex frequency is greater than or equal to about 30 megahertz. In a fourth embodiment of the RF duplex frequency, the RF duplex frequency is greater than or equal to about 40 megahertz. In a fifth embodiment of the RF duplex frequency, the RF duplex frequency is greater than or equal to about 50 megahertz.
- the PA bias circuitry 42 provides a PA bias signal PAB to the RF PA 40.
- the PA bias circuitry 42 biases the RF PA 40 via the PA bias signal PAB.
- the PA bias circuitry 42 biases the RF PA 40 based on the transmitter configuration signal PACS.
- the RF front-end circuitry 32 includes at least one RF switch, at least one RF amplifier, at least one RF filter, at least one RF duplexer, at least one RF diplexer, the like, or any combination thereof.
- the RF system control circuitry 30 is RF transceiver circuitry, which may include an RF transceiver IC, baseband controller circuitry, the like, or any combination thereof.
- the DC-DC converter 10 provides the first envelope power supply signal EPS, which has switching ripple.
- the first envelope power supply signal EPS provides power for amplification and at least partially envelope tracks the RF transmit signal RFT.
- Figure 7 shows the RF communications system 26 according to an alternate embodiment of the RF communications system 26.
- the RF transmitter circuitry 28 further includes a digital communications interface 44, which is coupled between the transmitter control circuitry 38 and a digital communications bus 46.
- the digital communications bus 46 is also coupled to the RF system control circuitry 30.
- the RF system control circuitry 30 provides the power supply control signal VRMP ( Figure 6) and the transmitter configuration signal PACS ( Figure 6) to the transmitter control circuitry 38 via the digital communications bus 46 and the digital communications interface 44.
- FIG 8 shows details of the DC-DC converter 10 illustrated in Figure 6 according to one embodiment of the DC-DC converter 10.
- the DC-DC converter 10 includes the power supply control circuitry 20, the parallel amplifier 14, and the switching supply 12.
- the power supply control circuitry 20 controls the parallel amplifier 14 and the switching supply 12.
- the parallel amplifier 14 and the switching supply 12 provide the first power supply output signal PS1 , such that the parallel amplifier 14 partially provides the first power supply output signal PS1 and the switching supply 12 partially provides the first power supply output signal PS1 .
- Figure 9 shows the RF communications system 26 according to another embodiment of the RF communications system 26.
- the PA bias circuitry 42 ( Figure 6) is omitted and the RF PA 40 includes a driver stage 48 and a final stage 50, which is coupled to the driver stage 48.
- the DC-DC converter 10 provides the second power supply output signal PS2, which is a second envelope power supply signal, to the driver stage 48 based on the power supply control signal VRMP. Further, the DC-DC converter 10 provides the first power supply output signal PS1 , which is the first envelope power supply signal, to the final stage 50 based on the power supply control signal VRMP.
- the driver stage 48 receives and amplifies the RF input signal RFI to provide a driver stage output signal DSO using the second envelope power supply signal, which provides power for amplification.
- the final stage 50 receives and amplifies the driver stage output signal DSO to provide the RF transmit signal RFT using the first envelope power supply signal, which provides power for amplification.
- Figure 10 shows the DC-DC converter 10 according to one
- the DC-DC converter 10 illustrated in Figure 10 is similar to the DC-DC converter 10 illustrated in Figure 1 , except in the DC-DC converter 10 illustrated in Figure 10, the switching supply 12 further includes a second inductive element L2.
- the second inductive element L2 is coupled between the feedback input FBI and the parallel amplifier output PAO.
- the switching supply 12 partially provides the first power supply output signal PS1 via the first inductive element L1 and the second inductive element L2.
- the switching supply 12 partially provides the first power supply output signal PS1 via a series combination of the first inductive element L1 and the second inductive element L2.
- a connection node 52 is provided where the first inductive element L1 and the second inductive element L2 are connected to one another.
- the connection node 52 provides a second voltage V2 to the parallel amplifier 14 via the feedback input FBI.
- the parallel amplifier 14 has a limited open loop gain at high frequencies that are above a frequency threshold. At such frequencies, a group delay in the parallel amplifier 14 may normally limit the ability of the parallel amplifier 14 to accurately regulate the first voltage V1 of the first power supply output signal PS1 .
- the parallel amplifier 14 partially provides the first power supply output signal PS1 via the parallel amplifier output PAO based on the voltage setpoint and feeding back a voltage to the feedback input FBI from the connection node 52 between the first inductive element L1 and the second inductive element L2.
- FIG. 1 1 shows the DC-DC converter 10 according to an alternate embodiment of the DC-DC converter 10.
- the DC-DC converter 10 illustrated in Figure 1 1 is similar to the DC-DC converter 10 illustrated in Figure 10, except the DC-DC converter 10 illustrated in Figure 1 1 further includes the offset capacitive element CO and the switching supply 12 further includes the filter capacitive element CF.
- the offset capacitive element CO is coupled between the parallel amplifier output PAO and the second inductive element L2.
- the parallel amplifier 14 partially provides the first power supply output signal PS1 via the parallel amplifier output PAO and the offset capacitive element CO based on the voltage setpoint.
- the first inductive element L1 and the second inductive element L2 provide a second power supply output signal PS2 via the connection node 52.
- the first inductive element L1 , the second inductive element L2, and the filter capacitive element CF form a first low-pass filter 54 having a first cutoff frequency.
- the second inductive element L2 and the filter capacitive element CF form a second low-pass filter 56 having a second cutoff frequency.
- the second cutoff frequency may be significantly higher than the first cutoff frequency.
- the first low-pass filter 54 may be used primarily to filter the switching output voltage VS, which is typically a square wave.
- the second low- pass filter 56 may be used to target specific high frequencies, such as certain harmonics of the switching output voltage VS.
- the second cutoff frequency is at least 10 times greater than the first cutoff frequency. In a second embodiment of the first low-pass filter 54 and the second low-pass filter 56, the second cutoff frequency is at least 100 times greater than the first cutoff frequency. In a third embodiment of the first low-pass filter 54 and the second low-pass filter 56, the second cutoff frequency is at least 500 times greater than the first cutoff frequency. In a fourth embodiment of the first low-pass filter 54 and the second low-pass filter 56, the second cutoff frequency is at least 1000 times greater than the first cutoff frequency.
- the second cutoff frequency is less than 1000 times greater than the first cutoff frequency. In a sixth embodiment of the first low-pass filter 54 and the second low-pass filter 56, the second cutoff frequency is less than 5000 times greater than the first cutoff frequency.
- the first inductive element L1 has a first inductance and the second inductive element L2 has a second inductance.
- a magnitude of the first inductance is at least 10 times greater than a magnitude of the second inductance.
- a magnitude of the first inductance is at least 100 times greater than a magnitude of the second inductance.
- a magnitude of the first inductance is at least 500 times greater than a magnitude of the second inductance.
- a magnitude of the first inductance is at least 1000 times greater than a magnitude of the second inductance.
- a magnitude of the first inductance is less than 1000 times greater than a magnitude of the second inductance.
- a magnitude of the first inductance is less than 5000 times greater than a magnitude of the second inductance.
- the first power supply output signal PS1 is fed to a load (not shown) having a load resistance RL, such as the RF PA 40 ( Figure 6).
- the switching output voltage VS has a DC component called a DC voltage VD and a ripple component called an AC voltage VA given by EQ. 1 , as shown below.
- the inductor current IL has a DC current I D and an AC current IA given by EQ. 2, as shown below.
- the DC-DC converter 10 regulates the DC voltage VD to be about equal to the voltage setpoint.
- the first inductive element L1 and the second inductive element L2 appear approximately as short circuits to the DC
- the filter capacitive element CF appears approximately as an open circuit to the DC component. Therefore, the DC voltage VD is approximately applied to the load resistance RL, as intended. As a result, the DC current ID is based on the DC voltage VD and the load resistance RL, as shown in EQ. 3 below.
- ID VD/RL.
- the first voltage V1 has a first residual ripple voltage VR1 and the second voltage V2 has a second residual ripple voltage VR2.
- the DC-DC converter 10 is the DC-DC converter 10 illustrated in Figure 10, such that the second voltage V2 is fed to the feedback input FBI, as shown.
- the second residual ripple voltage VR2 drives the parallel amplifier 14 to provide a ripple cancellation current, which is the parallel amplifier output current IP.
- the DC-DC converter 10 is similar to the DC-DC converter 10 illustrated in Figure 10, except the first voltage V1 is fed to the feedback input FBI instead of the second voltage V2, such that the first residual ripple voltage VR1 drives the parallel amplifier 14 to provide the ripple
- the open loop bandwidth factor T is small compared to one, such that the gain G approaches the DC open loop gain GO. Conversely, at frequencies significantly above the open loop bandwidth of the parallel amplifier 14, the open loop bandwidth factor T is large compared to one, such that the gain G approaches GO/sT.
- the parallel amplifier output current IP is based on the second residual ripple voltage VR2, as shown in EQ. 6 below.
- IP G * VR2 * (GO * VR2)/sT.
- the parallel amplifier output current IP is based on the first residual ripple voltage VR1 , as shown in EQ. 7 below.
- IP G * VR1 ⁇ (GO * VR1 )/sT.
- a difference between the first residual ripple voltage VR1 and the second residual ripple voltage VR2 is based on the AC current IA and the second inductance I2, as shown in EQ. 8 and EQ. 9 below.
- EQ. 1 1 IP * (GO)(VR1 )/sT + (GO)(IA)(l2)/T.
- EQ. 1 1 is representative of the first approach and EQ. 7 is representative of the second approach.
- the second residual ripple voltage VR2 drives the parallel amplifier 14 and in the second approach, the first residual ripple voltage VR1 drives the parallel amplifier 14.
- a smaller first residual ripple voltage VR1 represents better ripple cancellation performance.
- both approaches are assumed to provide the same magnitude of parallel amplifier output current IP.
- the parallel amplifier output current IP is phase-shifted from the first residual ripple voltage VR1 by about 90 degrees.
- the parallel amplifier output current IP is phase-shifted from the ripple current it is trying to cancel by about 90 degrees, thereby degrading ripple cancellation performance.
- the parallel amplifier output current IP has two terms, namely the
- the (GO)(VR1 )/sT term has the same phase-alignment shortcoming as in the second approach. But the
- (G0)(IA)(I2)/T term phase-aligns the parallel amplifier output current IP with the ripple current it is trying to cancel. Overall, the phase-alignment in the first approach is improved over the second approach. Additionally, to the extent that the (GO)(VR1 )/sT term is smaller than the (GO)(IA)(l2)/T term, the first residual ripple voltage VR1 is reduced, thereby improving ripple cancellation. In this regard, if the (GO)(IA)(l2)/T term is equal to the (GO)(VR1 )/sT term in EQ. 7, then in the (GO)(VR1 )/sT term in EQ. 1 1 , the first residual ripple voltage VR1 is equal to about zero, such that the first approach is greatly improved over the second approach.
- Figure 12 shows the RF communications system 26 according to one embodiment of the RF communications system 26.
- the RF communications system 26 illustrated in Figure 12 is similar to the RF communications system 26 illustrated in Figure 6, except in the RF communications system 26 illustrated in Figure 12 the transmitter control circuitry 38 provides a filter control signal FCS to the DC-DC converter 10.
- the transmitter control circuitry 38 selects the selectable notch frequency based on the RF duplex frequency.
- the filter control signal FCS is indicative of the selection of the selectable notch frequency.
- Figure 13 shows details of the DC-DC converter 10 illustrated in Figure 12 according to one embodiment of the DC-DC converter 10.
- the DC-DC converter 10 illustrated in Figure 13 is similar to the DC-DC converter 10 illustrated in Figure 1 , except in the DC-DC converter 10 illustrated in Figure 13, details of the switching supply 12 are not shown and the RF notch filter 18 receives the filter control signal FCS.
- FIG 14 shows details of the RF notch filter 18 used in the DC-DC converter 10 illustrated in Figure 13 according to one embodiment of the RF notch filter 18.
- the RF notch filter 18 includes a notch filter inductive element LT, a first notch filter capacitive element CT1 , a second notch filter capacitive element CT2, and a first switching element 58.
- the notch filter inductive element LT and the second notch filter capacitive element CT2 are coupled in series between the parallel amplifier output PAO and the ground.
- the first notch filter capacitive element CT1 and the first switching element 58 are coupled in series to form a series coupling, which is coupled across the second notch filter capacitive element CT2.
- a control input to the first switching element 58 receives the filter control signal FCS.
- the first switching element 58 is in one of an ON state and an OFF state based on the filter control signal FCS. Therefore, the selectable notch frequency is one of a first frequency and a second frequency.
- the selectable notch frequency is based on a combination of the notch filter inductive element LT in series with a parallel combination of the first notch filter capacitive element CT1 and the second notch filter capacitive element CT2.
- the selectable notch frequency is the first frequency.
- selectable notch frequency is based on a series combination of the notch filter inductive element LT and the second notch filter capacitive element CT2. As such, when the first switching element 58 is in the OFF state, the selectable notch frequency is the second frequency.
- the first frequency is equal to about 30 megahertz and the second frequency is equal to about 45
- the filter control signal FCS illustrated in Figure 14 is a single-bit signal, which minimizes control signal complexity. Further, in one embodiment of the RF notch filter 18, the notch filter inductive element LT and the second notch filter capacitive element CT2 are coupled directly in series between the parallel amplifier output PAO and the ground, which maximizes efficiency.
- FIG 15 shows the details of the RF notch filter 18 used in the DC- DC converter 10 illustrated in Figure 13 according to an alternate embodiment of the RF notch filter 18.
- the RF notch filter 18 includes the notch filter inductive element LT, the first notch filter capacitive element CT1 , the second notch filter capacitive element CT2, the first switching element 58, and a second switching element 60.
- the notch filter inductive element LT, the second notch filter capacitive element CT2, and the second switching element 60 are coupled in series between the parallel amplifier output PAO and the ground.
- the first notch filter capacitive element CT1 and the first switching element 58 are coupled in series to form a series coupling, which is coupled across a series combination of the second notch filter capacitive element CT2 and the second switching element 60.
- a control input to the first switching element 58 receives one bit of the filter control signal FCS.
- a control input to the second switching element 60 receives another bit of the filter control signal FCS.
- the first switching element 58 is in one of the ON state and the OFF state based on the filter control signal FCS
- the second switching element 60 is in one of an ON state and an OFF state based on the filter control signal FCS. Therefore, the selectable notch frequency is one of a first frequency, a second frequency, and a third frequency.
- the selectable notch frequency is based on a combination of the notch filter inductive element LT in series with a parallel combination of the first notch filter capacitive element CT1 and the second notch filter capacitive element CT2. As such, when the first switching element 58 is in the ON state and the second switching element 60 is in the ON state, the selectable notch frequency is the first frequency.
- the selectable notch frequency is based on a series combination of the notch filter inductive element LT and the second notch filter capacitive element CT2.
- the selectable notch frequency is the second frequency.
- the selectable notch frequency is based on a series combination of the notch filter inductive element LT and the first notch filter capacitive element CT1 .
- the selectable notch frequency is the third frequency.
- the first frequency is equal to about 30 megahertz
- the second frequency is equal to about 39 megahertz
- the third frequency is equal to about 47 megahertz.
- FIG. 1 6 shows the details of the RF notch filter 18 used in the DC- DC converter 10 illustrated in Figure 13 according to an additional embodiment of the RF notch filter 18.
- the RF notch filter 18 includes the notch filter inductive element LT, the first notch filter capacitive element CT1 and up to and including an N TH notch filter capacitive element CTN.
- the RF notch filter 18 further includes the first switching element 58 and up to and including an NTM switching element 62.
- the first notch filter capacitive element CT1 and the first switching element 58 are coupled in series to form a first series coupling.
- the N TH notch filter capacitive element CTN and the N TH switching element 62 are coupled in series to form an Nth series coupling.
- the RF notch filter 18 includes the first series coupling and up to and including the Nth series coupling to form a group of series couplings, such that each of the group of series couplings is coupled in parallel with one another.
- the notch filter inductive element LT and the group of series couplings are coupled in series between the parallel amplifier output PAO and the ground.
- Figure 17 is a graph illustrating frequency behavior of the RF
- the communications system 26 illustrated in Figure 12 according to one embodiment of the RF communications system 26.
- the graph illustrated in Figure 17 shows an RF spectrum associated with the RF transmit signal RFT ( Figure 12) and the RF receive signal RFR ( Figure 12)
- the RF receive signal RFR ( Figure 12) has an RF receive frequency FRX.
- the RF notch frequency RNF ( Figure 3) which is the selectable notch frequency
- the selectable notch frequency is selected to reduce noise at the RF receive frequency FRX.
- the RF transmit signal RFT ( Figure 12) is associated with a maximum transmit band 64.
- the RF transmit signal RFT ( Figure 12) is associated with a selected transmit band 66, which is a subset of the maximum transmit band 64.
- the maximum transmit band 64 has a maximum transmit bandwidth MTXB and the selected transmit band 66 has a selected transmit bandwidth STXB.
- the selected transmit bandwidth STXB is less than the maximum transmit bandwidth MTXB.
- the selected transmit band 66 may be used when the maximum transmit bandwidth MTXB is not required.
- the maximum transmit band 64 has a nominal RF transmit frequency FTXN, which is in the middle of the maximum transmit band 64.
- the selected transmit band 66 has a selected RF transmit frequency FTXS, which is in the middle of the selected transmit band 66.
- a selected transmit start offset STXO identifies the bottom edge of the selected transmit band 66 relative to the bottom edge of the maximum transmit band 64.
- An RF duplex frequency FDP is about equal to a difference between the RF receive frequency FRX and the nominal RF transmit frequency FTXN.
- the selectable notch frequency is selected to minimize a difference between the selectable notch frequency and the RF duplex frequency FDP, which may reduce noise at the RF receive frequency FRX.
- a preferred notch frequency FPN is about equal to a difference between the RF receive frequency FRX and the selected RF transmit frequency FTXS.
- the selectable notch frequency is selected to minimize a difference between the selectable notch frequency and the preferred notch frequency FPN, which may reduce noise at the RF receive frequency FRX.
- a duplex frequency correction FDC is about equal to a difference between the nominal RF transmit frequency FTXN and the selected RF transmit frequency FTXS. As such, the duplex frequency correction FDC is about equal to a difference between the preferred notch frequency FPN and the RF duplex frequency FDP. In one embodiment of the selected RF transmit frequency
- the selected RF transmit frequency FTXS is greater than the nominal RF transmit frequency FTXN, as shown in Figure 17. However, in an alternate embodiment of the selected RF transmit frequency FTXS, the selected RF transmit frequency FTXS is less than the nominal RF transmit frequency FTXN. Therefore, the duplex frequency correction FDC may be positive or negative.
- circuitry may use discrete circuitry, integrated circuitry, programmable circuitry, non-volatile circuitry, volatile circuitry, software executing instructions on computing hardware, firmware executing instructions on computing hardware, the like, or any combination thereof.
- the computing hardware may include mainframes, micro-processors, microcontrollers, DSPs, the like, or any combination thereof.
Abstract
A parallel amplifier (14), a switching supply (12), and a radio frequency (RF) notch filter (18) are disclosed. The parallel amplifier has a parallel amplifier output, such that the switching supply is coupled to the parallel amplifier output. Further, the RF notch filter is coupled between the parallel amplifier output and a ground. The RF notch filter has a selectable notch frequency, which is based on an RF duplex frequency.
Description
PROGRAMMABLE RF NOTCH FILTER FOR ENVELOPE TRACKING
Related Applications
[0001] This application claims the benefit of U.S. provisional patent
application number 61 /675,898, filed July 26, 2012, the disclosure of which is incorporated herein by reference in its entirety.
Field of the Disclosure
[0002] The present disclosure relates to direct current (DC)-DC converters and circuits that use DC-DC converters.
Background
[0003] DC-DC converters often include switching power supplies, which may be based on switching at least one end of an energy storage element, such as an inductor, between a source of DC voltage and a ground. As a result, an output voltage from a DC-DC converter may have a ripple voltage resulting from the switching associated with the energy storage element. Typically, the ripple voltage is undesirable and is minimized as much as sizes and costs permit.
Thus, there is a need to minimize ripple voltage using techniques that minimize sizes and costs.
Summary
[0004] A parallel amplifier, a switching supply, and a radio frequency (RF) notch filter are disclosed. The parallel amplifier has a parallel amplifier output, such that the switching supply is coupled to the parallel amplifier output. Further, the RF notch filter is coupled between the parallel amplifier output and a ground. The RF notch filter has a selectable notch frequency, which is based on an RF duplex frequency.
[0005] In one embodiment of the present disclosure, the parallel amplifier and the switching supply combine to provide a first envelope power supply signal to an RF power amplifier (PA), such that the first envelope power supply signal at
least partially envelope tracks an RF transmit signal, which is provided by the RF PA. In one embodiment of the RF notch filter, the RF notch filter reduces noise introduced by the first envelope power supply signal into RF receive circuitry. As such, the selectable notch frequency is selected to target specific receive frequencies.
[0006] Those skilled in the art will appreciate the scope of the disclosure and realize additional aspects thereof after reading the following detailed description in association with the accompanying drawings. Brief Description of the Drawings
[0007] The accompanying drawings incorporated in and forming a part of this specification illustrate several aspects of the disclosure, and together with the description serve to explain the principles of the disclosure.
[0008] Figure 1 shows a direct current (DC)-DC converter according to one embodiment of the present disclosure.
[0009] Figure 2 shows the DC-DC converter according to an alternate embodiment of the DC-DC converter.
[0010] Figure 3 is a graph illustrating a frequency response of a radio frequency (RF) notch filter illustrated in Figure 1 according to an additional embodiment of the DC-DC converter.
[0011] Figure 4 shows the DC-DC converter according to another
embodiment of the DC-DC converter.
[0012] Figure 5 shows the DC-DC converter according to a further
embodiment of the DC-DC converter.
[0013] Figure 6 shows a radio frequency (RF) communications system according to one embodiment of the present disclosure.
[0014] Figure 7 shows the RF communications system according to an alternate embodiment of the RF communications system.
[0015] Figure 8 shows the RF communications system according to an additional embodiment of the RF communications system.
[0016] Figure 9 shows the RF communications system according to another embodiment of the RF communications system.
[0017] Figure 10 shows the DC-DC converter according to one embodiment of the DC-DC converter.
[0018] Figure 1 1 shows the DC-DC converter according to an alternate embodiment of the DC-DC converter.
[0019] Figure 12 shows the RF communications system according to one embodiment of the RF communications system.
[0020] Figure 13 shows details of the DC-DC converter illustrated in Figure 12 according to one embodiment of the DC-DC converter.
[0021] Figure 14 shows details of an RF notch filter used in the DC-DC converter illustrated in Figure 13 according to one embodiment of the RF notch filter.
[0022] Figure 15 shows the details of the RF notch filter used in the DC-DC converter illustrated in Figure 13 according to an alternate embodiment of the RF notch filter.
[0023] Figure 1 6 shows the details of the RF notch filter used in the DC-DC converter illustrated in Figure 13 according to an additional embodiment of the RF notch filter.
[0024] Figure 17 is a graph illustrating frequency behavior of the RF communications system illustrated in Figure 12 according to one embodiment of the RF communications system.
Detailed Description
[0025] The embodiments set forth below represent the necessary information to enable those skilled in the art to practice the disclosure and illustrate the best mode of practicing the disclosure. Upon reading the following description in light of the accompanying drawings, those skilled in the art will understand the concepts of the disclosure and will recognize applications of these concepts not particularly addressed herein. It should be understood that these concepts and applications fall within the scope of the disclosure and the accompanying claims.
[0026] Figure 1 shows a direct current (DC)-DC converter 10 according to one embodiment of the present disclosure. The DC-DC converter 10 includes a switching supply 12, a parallel amplifier 14, and a radio frequency (RF) notch filter 18. The switching supply 12 includes switching circuitry 1 6 and a first inductive element L1 . The parallel amplifier 14 has a feedback input FBI and a parallel amplifier output PAO. In general, the switching supply 12 is coupled to the parallel amplifier output PAO. The switching circuitry 1 6 has a switching circuitry output SCO. The first inductive element L1 is coupled between the switching circuitry output SCO and the feedback input FBI. The RF notch filter 18 is coupled between the parallel amplifier output PAO and a ground. In one embodiment of the DC-DC converter 10, the parallel amplifier output PAO is directly coupled to the feedback input FBI, as shown.
[0027] In one embodiment of the DC-DC converter 10, the parallel amplifier 14 partially provides a first power supply output signal PS1 via the parallel amplifier output PAO based on a voltage setpoint. The switching supply 12 partially provides the first power supply output signal PS1 via the first inductive element L1 . The switching supply 12 may provide power more efficiently than the parallel amplifier 14. However, the parallel amplifier 14 may provide a voltage of the first power supply output signal PS1 more accurately than the switching supply 12. As such, in one embodiment of the DC-DC converter 10, the parallel amplifier 14 regulates the voltage, called a first voltage V1 , of the first power supply output signal PS1 based on the voltage setpoint of the first power supply output signal PS1 . Further, the switching supply 12 regulates the first power supply output signal PS1 to minimize an output current, called a parallel amplifier output current IP, from the parallel amplifier 14 to maximize efficiency. In this regard, the parallel amplifier 14 behaves like a voltage source and the switching supply 12 behaves like a current source. Additionally, the switching circuitry 1 6 provides a switching output voltage VS and an inductor current IL to the first inductive element L1 via the switching circuitry output SCO.
[0028] In one embodiment of the DC-DC converter 10, the DC-DC converter 10 receives a DC source signal VDC, such that the parallel amplifier 14 partially
provides the first power supply output signal PS1 using the DC source signal VDC and the switching supply 12 partially provides the first power supply output signal PS1 using the DC source signal VDC.
[0029] Figure 2 shows the DC-DC converter 10 according to an alternate embodiment of the DC-DC converter 10. The DC-DC converter 10 illustrated in Figure 2 is similar to the DC-DC converter 10 illustrated in Figure 1 , except the DC-DC converter 10 illustrated in Figure 2 further includes power supply control circuitry 20 and the switching supply 12 further includes a filter capacitive element CF. The filter capacitive element CF is coupled between the parallel amplifier output PAO and the ground. As such, the filter capacitive element CF may significantly reduce unwanted noise, ripple, or both from the first power supply output signal PS1 . The power supply control circuitry 20 receives the DC source signal VDC and is coupled to the parallel amplifier 14 and the switching circuitry 1 6.
[0030] Figure 3 is a graph illustrating a frequency response 22 of the RF notch filter 18 (Figure 1 ) illustrated in Figure 1 according to an additional embodiment of the DC-DC converter 10. The RF notch filter 18 (Figure 1 ) has the frequency response 22 with an RF notch 24 at an RF notch frequency RNF. Therefore, the RF notch filter 18 (Figure 1 ) filters the first power supply output signal PS1 (Figure 1 ) based on the frequency response 22. As such, the RF notch filter 18 (Figure 1 ) may significantly reduce unwanted noise, ripple, or both from the first power supply output signal PS1 (Figure 1 ) at the RF notch frequency RNF. In one embodiment of the RF notch filter 18 (Figure 1 ), the RF notch filter 18 (Figure 1 ) is a programmable RF notch filter, such that the RF notch frequency RNF is a selectable notch frequency. In an alternate
embodiment of the RF notch filter 18 (Figure 1 ), the RF notch filter 18 (Figure 1 ) is a fixed RF notch filter, such that the RF notch frequency RNF is not a selectable notch frequency.
[0031] In a first embodiment of the frequency response 22, the RF notch frequency RNF is equal to about 10 megahertz. In a second embodiment of the frequency response 22, the RF notch frequency RNF is equal to about 20
megahertz. In a third embodiment of the frequency response 22, the RF notch frequency RNF is equal to about 30 megahertz. In a fourth embodiment of the frequency response 22, the RF notch frequency RNF is equal to about 40 megahertz. In a fifth embodiment of the frequency response 22, the RF notch frequency RNF is equal to about 50 megahertz.
[0032] Figure 4 shows the DC-DC converter 10 according to another embodiment of the DC-DC converter 10. The DC-DC converter 10 illustrated in Figure 4 is similar to the DC-DC converter 10 illustrated in Figure 2, except the DC-DC converter 10 illustrated in Figure 4 further includes an offset capacitive element CO coupled between the parallel amplifier output PAO and the feedback input FBI. As such, the switching supply 12 is coupled to the parallel amplifier output PAO via the offset capacitive element CO and the RF notch filter 18 is coupled to the parallel amplifier output PAO via the offset capacitive element CO. Additionally, the RF notch filter 18 includes a notch filter capacitive element CT and a notch filter inductive element LT coupled in series. The notch filter capacitive element CT and the notch filter inductive element LT form a resonant circuit having a resonant frequency. The RF notch frequency RNF (Figure 3) is based on the resonant frequency. A shape of the frequency response 22 (Figure 3) near the RF notch frequency RNF (Figure 3) may be based on a Q factor of the resonant circuit.
[0033] The parallel amplifier 14 partially provides the first power supply output signal PS1 via the parallel amplifier output PAO and the offset capacitive element CO based on the voltage setpoint. The offset capacitive element CO allows the first voltage V1 to be higher than a voltage at the parallel amplifier output PAO. As a result, the parallel amplifier 14 may properly regulate the first voltage V1 even if the first voltage V1 is greater than a maximum output voltage from the parallel amplifier 14 at the parallel amplifier output PAO. In the embodiment of the DC-DC converter 10 illustrated in Figure 4, the filter capacitive element CF is coupled between the parallel amplifier output PAO and the ground through the offset capacitive element CO. In an alternate embodiment of the DC-DC converter 10, the offset capacitive element CO is omitted.
[0034] Figure 5 shows the DC-DC converter 10 according to a further embodiment of the DC-DC converter 10. The DC-DC converter 10 illustrated in Figure 5 is similar to the DC-DC converter 10 illustrated in Figure 4, except in the DC-DC converter 10 illustrated in Figure 5, the RF notch filter 18 includes the notch filter capacitive element CT, the notch filter inductive element LT, and a notch filter resistive element RT coupled in series. The notch filter capacitive element CT, the notch filter inductive element LT, and the notch filter resistive element RT form a resonant circuit having a resonant frequency. The RF notch frequency RNF (Figure 3) is based on the resonant frequency. A shape of the frequency response 22 (Figure 3) near the RF notch frequency RNF (Figure 3) may be based on a Q factor of the resonant circuit.
[0035] Figure 6 shows a radio frequency (RF) communications system 26 according to one embodiment of the present disclosure. The RF
communications system 26 includes RF transmitter circuitry 28, RF system control circuitry 30, RF front-end circuitry 32, an RF antenna 34, and a DC power source 36. The RF transmitter circuitry 28 includes transmitter control circuitry 38, an RF power amplifier (PA) 40, the DC-DC converter 10, and PA bias circuitry 42. The DC-DC converter 10 functions as an envelope tracking power supply. In an alternate embodiment of the RF communications system 26, the DC power source 36 is external to the RF communications system 26.
[0036] In one embodiment of the RF communications system 26, the RF front- end circuitry 32 receives via the RF antenna 34, processes, and forwards an RF receive signal RFR to the RF system control circuitry 30. In one embodiment of the RF communications system 26, the RF receive signal RFR has an RF receive frequency. Further, the RF notch frequency RNF (Figure 3) is about equal to the RF receive frequency, which may reduce noise, ripple, or both in the receive path from the transmit path or other noise sources. The RF system control circuitry 30 provides a power supply control signal VRMP and a
transmitter configuration signal PACS to the transmitter control circuitry 38. The RF system control circuitry 30 provides an RF input signal RFI to the RF PA 40. The DC power source 36 provides a DC source signal VDC to the DC-DC
converter 10. In one embodiment of the DC power source 36, the DC power source 36 is a battery. In one embodiment of the power supply control signal VRMP, the power supply control signal VRMP is an envelope power supply control signal. Specifically, the DC power source 36 provides the DC source signal VDC to the parallel amplifier 14 (Figure 1 ) and to the switching supply 12 (Figure 1 ).
[0037] The transmitter control circuitry 38 is coupled to the DC-DC converter 10 and to the PA bias circuitry 42. The DC-DC converter 10 provides the first power supply output signal PS1 to the RF PA 40 based on the power supply control signal VRMP. In this regard, the DC-DC converter 10 is an envelope tracking power supply and the first power supply output signal PS1 is a first envelope power supply signal EPS. The DC source signal VDC provides power to the DC-DC converter 10. As such, the first power supply output signal PS1 , which is the first envelope power supply signal EPS, is based on the DC source signal VDC. The power supply control signal VRMP is representative of a voltage setpoint of the first envelope power supply signal EPS. The RF PA 40 receives and amplifies the RF input signal RFI to provide an RF transmit signal RFT using the first envelope power supply signal EPS. The first envelope power supply signal EPS provides power for amplification to the RF PA 40.
[0038] In one embodiment of the DC-DC converter 10, the first envelope power supply signal EPS is amplitude modulated to at least partially provide envelope tracking. In one embodiment of the RF PA 40, the RF PA 40 operates with approximately constant gain, called isogain, and with gain compression. In a first embodiment of the gain compression, the gain compression is greater than about one decibel. In a second embodiment of the gain compression, the gain compression is greater than about two decibels. In a third embodiment of the gain compression, the gain compression is equal to about two decibels. In a fourth embodiment of the gain compression, the gain compression is equal to about three decibels. In a fifth embodiment of the gain compression, the gain compression is equal to about four decibels. By operating with higher levels of
gain compression, efficiency of the RF PA 40 may be increased, which may help compensate for reduced efficiency in the DC-DC converter 10.
[0039] In a first embodiment of the first envelope power supply signal EPS, a bandwidth of the first envelope power supply signal EPS is greater than or equal to about 10 megahertz. In a second embodiment of the first envelope power supply signal EPS, a bandwidth of the first envelope power supply signal EPS is less than or equal to about 10 megahertz. In a third embodiment of the first envelope power supply signal EPS, a bandwidth of the first envelope power supply signal EPS is greater than or equal to about 20 megahertz. In a fourth embodiment of the first envelope power supply signal EPS, a bandwidth of the first envelope power supply signal EPS is less than or equal to about 20 megahertz.
[0040] The RF front-end circuitry 32 receives, processes, and transmits the RF transmit signal RFT via the RF antenna 34. In one embodiment of the RF transmitter circuitry 28, the transmitter control circuitry 38 configures the RF transmitter circuitry 28 based on the transmitter configuration signal PACS. In one embodiment of the RF communications system 26, the RF communications system 26 operates in a full duplex environment, such that the RF transmit signal RFT and the RF receive signal RFR may be active simultaneously. The RF transmit signal RFT has an RF transmit frequency and the RF receive signal RFR has the RF receive frequency. A difference between the RF transmit frequency and the RF receive frequency is about equal to an RF duplex frequency. In one embodiment of the RF communications system 26, the RF notch frequency RNF (Figure 3) is about equal to the RF duplex frequency, which may reduce noise in the receive path from the transmit path.
[0041] In one embodiment of the RF notch filter 18 (Figure 1 ), the RF notch filter 18 (Figure 1 ) is a programmable RF notch filter, such that the RF notch frequency RNF is a selectable notch frequency. In one embodiment of the programmable RF notch filter, the selectable notch frequency is based on the RF duplex frequency. In one embodiment of the programmable RF notch filter, the selectable notch frequency is about equal to the RF duplex frequency. In one
embodiment of the RF communications system 26, the transmitter control circuitry 38 selects the selectable notch frequency based on the RF duplex frequency. In one embodiment of the RF communications system 26, the RF system control circuitry 30 provides notch frequency information to the
transmitter control circuitry 38 via the transmitter configuration signal PACS. The notch frequency information is based on the RF duplex frequency. Then, the transmitter control circuitry 38 selects the selectable notch frequency based on the notch frequency information. As such, selection of the selectable notch frequency is based on the notch frequency information.
[0042] In a first embodiment of the RF duplex frequency, the RF duplex frequency is greater than or equal to about 10 megahertz. In a second embodiment of the RF duplex frequency, the RF duplex frequency is greater than or equal to about 20 megahertz. In a third embodiment of the RF duplex frequency, the RF duplex frequency is greater than or equal to about 30 megahertz. In a fourth embodiment of the RF duplex frequency, the RF duplex frequency is greater than or equal to about 40 megahertz. In a fifth embodiment of the RF duplex frequency, the RF duplex frequency is greater than or equal to about 50 megahertz.
[0043] The PA bias circuitry 42 provides a PA bias signal PAB to the RF PA 40. In this regard, the PA bias circuitry 42 biases the RF PA 40 via the PA bias signal PAB. In one embodiment of the PA bias circuitry 42, the PA bias circuitry 42 biases the RF PA 40 based on the transmitter configuration signal PACS. In one embodiment of the RF front-end circuitry 32, the RF front-end circuitry 32 includes at least one RF switch, at least one RF amplifier, at least one RF filter, at least one RF duplexer, at least one RF diplexer, the like, or any combination thereof. In one embodiment of the RF system control circuitry 30, the RF system control circuitry 30 is RF transceiver circuitry, which may include an RF transceiver IC, baseband controller circuitry, the like, or any combination thereof. In one embodiment of the RF transmitter circuitry 28, the DC-DC converter 10 provides the first envelope power supply signal EPS, which has switching ripple. In one embodiment of the RF transmitter circuitry 28, the first envelope power
supply signal EPS provides power for amplification and at least partially envelope tracks the RF transmit signal RFT.
[0044] Figure 7 shows the RF communications system 26 according to an alternate embodiment of the RF communications system 26. The RF
communications system 26 illustrated in Figure 7 is similar to the RF
communications system 26 illustrated in Figure 6, except in the RF
communications system 26 illustrated in Figure 7, the RF transmitter circuitry 28 further includes a digital communications interface 44, which is coupled between the transmitter control circuitry 38 and a digital communications bus 46. The digital communications bus 46 is also coupled to the RF system control circuitry 30. As such, the RF system control circuitry 30 provides the power supply control signal VRMP (Figure 6) and the transmitter configuration signal PACS (Figure 6) to the transmitter control circuitry 38 via the digital communications bus 46 and the digital communications interface 44.
[0045] Figure 8 shows details of the DC-DC converter 10 illustrated in Figure 6 according to one embodiment of the DC-DC converter 10. The DC-DC converter 10 includes the power supply control circuitry 20, the parallel amplifier 14, and the switching supply 12. The power supply control circuitry 20 controls the parallel amplifier 14 and the switching supply 12. The parallel amplifier 14 and the switching supply 12 provide the first power supply output signal PS1 , such that the parallel amplifier 14 partially provides the first power supply output signal PS1 and the switching supply 12 partially provides the first power supply output signal PS1 .
[0046] Figure 9 shows the RF communications system 26 according to another embodiment of the RF communications system 26. The RF
communications system 26 illustrated in Figure 9 is similar to the RF
communications system 26 illustrated in Figure 6, except in the RF
communications system 26 illustrated in Figure 9, the PA bias circuitry 42 (Figure 6) is omitted and the RF PA 40 includes a driver stage 48 and a final stage 50, which is coupled to the driver stage 48. The DC-DC converter 10 provides the second power supply output signal PS2, which is a second envelope power
supply signal, to the driver stage 48 based on the power supply control signal VRMP. Further, the DC-DC converter 10 provides the first power supply output signal PS1 , which is the first envelope power supply signal, to the final stage 50 based on the power supply control signal VRMP. The driver stage 48 receives and amplifies the RF input signal RFI to provide a driver stage output signal DSO using the second envelope power supply signal, which provides power for amplification. Similarly, the final stage 50 receives and amplifies the driver stage output signal DSO to provide the RF transmit signal RFT using the first envelope power supply signal, which provides power for amplification.
[0047] Figure 10 shows the DC-DC converter 10 according to one
embodiment of the DC-DC converter 10. The DC-DC converter 10 illustrated in Figure 10 is similar to the DC-DC converter 10 illustrated in Figure 1 , except in the DC-DC converter 10 illustrated in Figure 10, the switching supply 12 further includes a second inductive element L2. The second inductive element L2 is coupled between the feedback input FBI and the parallel amplifier output PAO. The switching supply 12 partially provides the first power supply output signal PS1 via the first inductive element L1 and the second inductive element L2. Specifically, the switching supply 12 partially provides the first power supply output signal PS1 via a series combination of the first inductive element L1 and the second inductive element L2.
[0048] In one embodiment of the switching supply 12, a connection node 52 is provided where the first inductive element L1 and the second inductive element L2 are connected to one another. The connection node 52 provides a second voltage V2 to the parallel amplifier 14 via the feedback input FBI. Further, in one embodiment of the parallel amplifier 14, the parallel amplifier 14 has a limited open loop gain at high frequencies that are above a frequency threshold. At such frequencies, a group delay in the parallel amplifier 14 may normally limit the ability of the parallel amplifier 14 to accurately regulate the first voltage V1 of the first power supply output signal PS1 . However, by feeding back the second voltage V2 to the feedback input FBI instead of the first voltage V1 , a phase-shift that is developed across the second inductive element L2 at least partially
compensates for the limited open loop gain of the parallel amplifier 14 at frequencies that are above the frequency threshold, thereby improving the ability of the parallel amplifier 14 to accurately regulate the first voltage V1 . In this regard, in one embodiment of the DC-DC converter 10, the parallel amplifier 14 partially provides the first power supply output signal PS1 via the parallel amplifier output PAO based on the voltage setpoint and feeding back a voltage to the feedback input FBI from the connection node 52 between the first inductive element L1 and the second inductive element L2.
[0049] Figure 1 1 shows the DC-DC converter 10 according to an alternate embodiment of the DC-DC converter 10. The DC-DC converter 10 illustrated in Figure 1 1 is similar to the DC-DC converter 10 illustrated in Figure 10, except the DC-DC converter 10 illustrated in Figure 1 1 further includes the offset capacitive element CO and the switching supply 12 further includes the filter capacitive element CF. The offset capacitive element CO is coupled between the parallel amplifier output PAO and the second inductive element L2. In one embodiment of the DC-DC converter 10, the parallel amplifier 14 partially provides the first power supply output signal PS1 via the parallel amplifier output PAO and the offset capacitive element CO based on the voltage setpoint. The first inductive element L1 and the second inductive element L2 provide a second power supply output signal PS2 via the connection node 52.
[0050] The first inductive element L1 , the second inductive element L2, and the filter capacitive element CF form a first low-pass filter 54 having a first cutoff frequency. The second inductive element L2 and the filter capacitive element CF form a second low-pass filter 56 having a second cutoff frequency. The second cutoff frequency may be significantly higher than the first cutoff frequency. As such, the first low-pass filter 54 may be used primarily to filter the switching output voltage VS, which is typically a square wave. However, the second low- pass filter 56 may be used to target specific high frequencies, such as certain harmonics of the switching output voltage VS.
[0051] In a first embodiment of the first low-pass filter 54 and the second low- pass filter 56, the second cutoff frequency is at least 10 times greater than the
first cutoff frequency. In a second embodiment of the first low-pass filter 54 and the second low-pass filter 56, the second cutoff frequency is at least 100 times greater than the first cutoff frequency. In a third embodiment of the first low-pass filter 54 and the second low-pass filter 56, the second cutoff frequency is at least 500 times greater than the first cutoff frequency. In a fourth embodiment of the first low-pass filter 54 and the second low-pass filter 56, the second cutoff frequency is at least 1000 times greater than the first cutoff frequency. In a fifth embodiment of the first low-pass filter 54 and the second low-pass filter 56, the second cutoff frequency is less than 1000 times greater than the first cutoff frequency. In a sixth embodiment of the first low-pass filter 54 and the second low-pass filter 56, the second cutoff frequency is less than 5000 times greater than the first cutoff frequency.
[0052] The first inductive element L1 has a first inductance and the second inductive element L2 has a second inductance. In a first embodiment of the first inductive element L1 and the second inductive element L2, a magnitude of the first inductance is at least 10 times greater than a magnitude of the second inductance. In a second embodiment of the first inductive element L1 and the second inductive element L2, a magnitude of the first inductance is at least 100 times greater than a magnitude of the second inductance. In a third embodiment of the first inductive element L1 and the second inductive element L2, a magnitude of the first inductance is at least 500 times greater than a magnitude of the second inductance. In a fourth embodiment of the first inductive element L1 and the second inductive element L2, a magnitude of the first inductance is at least 1000 times greater than a magnitude of the second inductance. In a fifth embodiment of the first inductive element L1 and the second inductive element L2, a magnitude of the first inductance is less than 1000 times greater than a magnitude of the second inductance. In a sixth embodiment of the first inductive element L1 and the second inductive element L2, a magnitude of the first inductance is less than 5000 times greater than a magnitude of the second inductance.
[0053] An analysis of improved ripple cancellation performance of the DC-DC converter 10 illustrated in Figure 1 1 is presented. In general, the first power supply output signal PS1 is fed to a load (not shown) having a load resistance RL, such as the RF PA 40 (Figure 6). The switching output voltage VS has a DC component called a DC voltage VD and a ripple component called an AC voltage VA given by EQ. 1 , as shown below.
EQ. 1 : VS = VD + VA. [0054] Further, the inductor current IL has a DC current I D and an AC current IA given by EQ. 2, as shown below.
EQ. 2: IL = ID + IA. [0055] The DC-DC converter 10 regulates the DC voltage VD to be about equal to the voltage setpoint. The first inductive element L1 and the second inductive element L2 appear approximately as short circuits to the DC
component. Further, the filter capacitive element CF appears approximately as an open circuit to the DC component. Therefore, the DC voltage VD is approximately applied to the load resistance RL, as intended. As a result, the DC current ID is based on the DC voltage VD and the load resistance RL, as shown in EQ. 3 below.
EQ. 3: ID = VD/RL.
[0056] Most of the ripple components of the switching output voltage VS is filtered out from the first voltage V1 by the first low-pass filter 54 and the second low-pass filter 56. As a result, most of the AC voltage VA is across the series combination of the first inductive element L1 and the second inductive element L2. The first inductive element L1 has a first inductance 11 and the second inductive element L2 has a second inductance I2. Therefore, the AC current IA is
based on the AC voltage VA, the first inductance 11 and the second inductance I2, where s=j2nf, j=V-1 , and f=frequency, as shown in EQ. 4 below.
EQ. 4: IA = VA/[S(I1 +I2)].
[0057] Much of what remains of the ripple component is cancelled out from the first voltage V1 by the parallel amplifier 14. However, to the extent that the parallel amplifier 14 cannot completely cancel out the remains of the ripple component, the first voltage V1 has a first residual ripple voltage VR1 and the second voltage V2 has a second residual ripple voltage VR2. Two approaches to ripple cancellation will be compared against one another. In the first approach, the DC-DC converter 10 is the DC-DC converter 10 illustrated in Figure 10, such that the second voltage V2 is fed to the feedback input FBI, as shown. In this regard, the second residual ripple voltage VR2 drives the parallel amplifier 14 to provide a ripple cancellation current, which is the parallel amplifier output current IP. In the second approach, the DC-DC converter 10 is similar to the DC-DC converter 10 illustrated in Figure 10, except the first voltage V1 is fed to the feedback input FBI instead of the second voltage V2, such that the first residual ripple voltage VR1 drives the parallel amplifier 14 to provide the ripple
cancellation current, which is the parallel amplifier output current IP.
[0058] In the following analysis, the parallel amplifier 14 has a DC open loop gain GO and an open loop bandwidth factor T. As a result, the parallel amplifier 14 has a gain G, as shown in EQ. 5 below. EQ. 5: G = GO/(1 +sT).
[0059] As a result, at frequencies significantly below an open loop bandwidth of the parallel amplifier 14, the open loop bandwidth factor T is small compared to one, such that the gain G approaches the DC open loop gain GO. Conversely, at frequencies significantly above the open loop bandwidth of the parallel
amplifier 14, the open loop bandwidth factor T is large compared to one, such that the gain G approaches GO/sT.
[0060] In the first approach, described above wherein the second residual ripple voltage VR2 drives the parallel amplifier 14 and at frequencies significantly above the open loop bandwidth of the parallel amplifier 14, the parallel amplifier output current IP is based on the second residual ripple voltage VR2, as shown in EQ. 6 below.
EQ. 6: IP = G * VR2 * (GO * VR2)/sT.
[0061 ] In the second approach described above, when the first residual ripple voltage VR1 drives the parallel amplifier 14 and at frequencies significantly above the open loop bandwidth of the parallel amplifier 14, the parallel amplifier output current IP is based on the first residual ripple voltage VR1 , as shown in EQ. 7 below.
EQ. 7: IP = G * VR1 ~ (GO * VR1 )/sT.
[0062] However, a difference between the first residual ripple voltage VR1 and the second residual ripple voltage VR2 is based on the AC current IA and the second inductance I2, as shown in EQ. 8 and EQ. 9 below.
EQ. 8: (VR2 - VR1 ) = (S)(IA)(I2), or
EQ. 9: VR2 = (s)(IA)(l2) + VR1 .
[0063] Substituting EQ. 9 into EQ. 6 provides EQ. 10 and EQ. 1 1 , as shown below.
EQ. 10: IP * (GO)(VR1 )/sT + (GO)(s)(IA)(l2)/sT, or
EQ. 1 1 : IP * (GO)(VR1 )/sT + (GO)(IA)(l2)/T.
[0064] EQ. 1 1 is representative of the first approach and EQ. 7 is representative of the second approach. As a reminder, in the first approach, the second residual ripple voltage VR2 drives the parallel amplifier 14 and in the second approach, the first residual ripple voltage VR1 drives the parallel amplifier 14. In both equations, a smaller first residual ripple voltage VR1 represents better ripple cancellation performance. For comparison purposes, both approaches are assumed to provide the same magnitude of parallel amplifier output current IP. However, in the second approach, the parallel amplifier output current IP is phase-shifted from the first residual ripple voltage VR1 by about 90 degrees. As such, the parallel amplifier output current IP is phase-shifted from the ripple current it is trying to cancel by about 90 degrees, thereby degrading ripple cancellation performance. However, in the first approach, according to EQ. 1 1 , the parallel amplifier output current IP has two terms, namely the
(GO)(VR1 )/sT term and the (G0)(IA)(I2)/T term. The (GO)(VR1 )/sT term has the same phase-alignment shortcoming as in the second approach. But the
(G0)(IA)(I2)/T term phase-aligns the parallel amplifier output current IP with the ripple current it is trying to cancel. Overall, the phase-alignment in the first approach is improved over the second approach. Additionally, to the extent that the (GO)(VR1 )/sT term is smaller than the (GO)(IA)(l2)/T term, the first residual ripple voltage VR1 is reduced, thereby improving ripple cancellation. In this regard, if the (GO)(IA)(l2)/T term is equal to the (GO)(VR1 )/sT term in EQ. 7, then in the (GO)(VR1 )/sT term in EQ. 1 1 , the first residual ripple voltage VR1 is equal to about zero, such that the first approach is greatly improved over the second approach.
[0065] Figure 12 shows the RF communications system 26 according to one embodiment of the RF communications system 26. The RF communications system 26 illustrated in Figure 12 is similar to the RF communications system 26 illustrated in Figure 6, except in the RF communications system 26 illustrated in Figure 12 the transmitter control circuitry 38 provides a filter control signal FCS to the DC-DC converter 10. In one embodiment of the RF communications system 26, the transmitter control circuitry 38 selects the selectable notch frequency
based on the RF duplex frequency. As such, the filter control signal FCS is indicative of the selection of the selectable notch frequency.
[0066] Figure 13 shows details of the DC-DC converter 10 illustrated in Figure 12 according to one embodiment of the DC-DC converter 10. The DC-DC converter 10 illustrated in Figure 13 is similar to the DC-DC converter 10 illustrated in Figure 1 , except in the DC-DC converter 10 illustrated in Figure 13, details of the switching supply 12 are not shown and the RF notch filter 18 receives the filter control signal FCS.
[0067] Figure 14 shows details of the RF notch filter 18 used in the DC-DC converter 10 illustrated in Figure 13 according to one embodiment of the RF notch filter 18. The RF notch filter 18 includes a notch filter inductive element LT, a first notch filter capacitive element CT1 , a second notch filter capacitive element CT2, and a first switching element 58. The notch filter inductive element LT and the second notch filter capacitive element CT2 are coupled in series between the parallel amplifier output PAO and the ground. The first notch filter capacitive element CT1 and the first switching element 58 are coupled in series to form a series coupling, which is coupled across the second notch filter capacitive element CT2.
[0068] A control input to the first switching element 58 receives the filter control signal FCS. As such, the first switching element 58 is in one of an ON state and an OFF state based on the filter control signal FCS. Therefore, the selectable notch frequency is one of a first frequency and a second frequency. When the first switching element 58 is in the ON state, the selectable notch frequency is based on a combination of the notch filter inductive element LT in series with a parallel combination of the first notch filter capacitive element CT1 and the second notch filter capacitive element CT2. As such, when the first switching element 58 is in the ON state, the selectable notch frequency is the first frequency. When the first switching element 58 is in the OFF state, the
selectable notch frequency is based on a series combination of the notch filter inductive element LT and the second notch filter capacitive element CT2. As
such, when the first switching element 58 is in the OFF state, the selectable notch frequency is the second frequency.
[0069] In one embodiment of the RF notch filter 18, the first frequency is equal to about 30 megahertz and the second frequency is equal to about 45
megahertz. The filter control signal FCS illustrated in Figure 14 is a single-bit signal, which minimizes control signal complexity. Further, in one embodiment of the RF notch filter 18, the notch filter inductive element LT and the second notch filter capacitive element CT2 are coupled directly in series between the parallel amplifier output PAO and the ground, which maximizes efficiency.
[0070] Figure 15 shows the details of the RF notch filter 18 used in the DC- DC converter 10 illustrated in Figure 13 according to an alternate embodiment of the RF notch filter 18. The RF notch filter 18 includes the notch filter inductive element LT, the first notch filter capacitive element CT1 , the second notch filter capacitive element CT2, the first switching element 58, and a second switching element 60. The notch filter inductive element LT, the second notch filter capacitive element CT2, and the second switching element 60 are coupled in series between the parallel amplifier output PAO and the ground. The first notch filter capacitive element CT1 and the first switching element 58 are coupled in series to form a series coupling, which is coupled across a series combination of the second notch filter capacitive element CT2 and the second switching element 60.
[0071] A control input to the first switching element 58 receives one bit of the filter control signal FCS. A control input to the second switching element 60 receives another bit of the filter control signal FCS. As such, the first switching element 58 is in one of the ON state and the OFF state based on the filter control signal FCS, and the second switching element 60 is in one of an ON state and an OFF state based on the filter control signal FCS. Therefore, the selectable notch frequency is one of a first frequency, a second frequency, and a third frequency. When the first switching element 58 is in the ON state and the second switching element 60 is in the ON state, the selectable notch frequency is based on a combination of the notch filter inductive element LT in series with a parallel
combination of the first notch filter capacitive element CT1 and the second notch filter capacitive element CT2. As such, when the first switching element 58 is in the ON state and the second switching element 60 is in the ON state, the selectable notch frequency is the first frequency.
[0072] When the first switching element 58 is in the OFF state and the second switching element 60 is in the ON state, the selectable notch frequency is based on a series combination of the notch filter inductive element LT and the second notch filter capacitive element CT2. As such, when the first switching element 58 is in the OFF state and the second switching element 60 is in the ON state, the selectable notch frequency is the second frequency. When the first switching element 58 is in the ON state and the second switching element 60 is in the OFF state, the selectable notch frequency is based on a series combination of the notch filter inductive element LT and the first notch filter capacitive element CT1 . As such, when the first switching element 58 is in the ON state and the second switching element 60 is in the OFF state, the selectable notch frequency is the third frequency.
[0073] In one embodiment of the RF notch filter 18, the first frequency is equal to about 30 megahertz, the second frequency is equal to about 39 megahertz, and the third frequency is equal to about 47 megahertz. When the first switching element 58 is in the OFF state and the second switching element 60 is in the OFF state, the RF notch filter 18 is disabled.
[0074] Figure 1 6 shows the details of the RF notch filter 18 used in the DC- DC converter 10 illustrated in Figure 13 according to an additional embodiment of the RF notch filter 18. The RF notch filter 18 includes the notch filter inductive element LT, the first notch filter capacitive element CT1 and up to and including an NTH notch filter capacitive element CTN. The RF notch filter 18 further includes the first switching element 58 and up to and including an N™ switching element 62. The first notch filter capacitive element CT1 and the first switching element 58 are coupled in series to form a first series coupling. The NTH notch filter capacitive element CTN and the NTH switching element 62 are coupled in series to form an Nth series coupling. As such, the RF notch filter 18 includes
the first series coupling and up to and including the Nth series coupling to form a group of series couplings, such that each of the group of series couplings is coupled in parallel with one another. As such, the notch filter inductive element LT and the group of series couplings are coupled in series between the parallel amplifier output PAO and the ground.
[0075] Figure 17 is a graph illustrating frequency behavior of the RF
communications system 26 illustrated in Figure 12 according to one embodiment of the RF communications system 26. The graph illustrated in Figure 17 shows an RF spectrum associated with the RF transmit signal RFT (Figure 12) and the RF receive signal RFR (Figure 12) The RF receive signal RFR (Figure 12) has an RF receive frequency FRX. In one embodiment of the RF notch frequency RNF (Figure 3), which is the selectable notch frequency, the selectable notch frequency is selected to reduce noise at the RF receive frequency FRX. In one embodiment of the RF transmit signal RFT (Figure 12), the RF transmit signal RFT (Figure 12) is associated with a maximum transmit band 64. In an alternate embodiment of the RF transmit signal RFT (Figure 12), the RF transmit signal RFT (Figure 12) is associated with a selected transmit band 66, which is a subset of the maximum transmit band 64.
[0076] The maximum transmit band 64 has a maximum transmit bandwidth MTXB and the selected transmit band 66 has a selected transmit bandwidth STXB. The selected transmit bandwidth STXB is less than the maximum transmit bandwidth MTXB. As such, the selected transmit band 66 may be used when the maximum transmit bandwidth MTXB is not required. The maximum transmit band 64 has a nominal RF transmit frequency FTXN, which is in the middle of the maximum transmit band 64. The selected transmit band 66 has a selected RF transmit frequency FTXS, which is in the middle of the selected transmit band 66. A selected transmit start offset STXO identifies the bottom edge of the selected transmit band 66 relative to the bottom edge of the maximum transmit band 64.
[0077] An RF duplex frequency FDP is about equal to a difference between the RF receive frequency FRX and the nominal RF transmit frequency FTXN. In
one embodiment of the RF notch frequency RNF (Figure 3), which is the selectable notch frequency, the selectable notch frequency is selected to minimize a difference between the selectable notch frequency and the RF duplex frequency FDP, which may reduce noise at the RF receive frequency FRX. A preferred notch frequency FPN is about equal to a difference between the RF receive frequency FRX and the selected RF transmit frequency FTXS. In one embodiment of the RF notch frequency RNF (Figure 3), which is the selectable notch frequency, the selectable notch frequency is selected to minimize a difference between the selectable notch frequency and the preferred notch frequency FPN, which may reduce noise at the RF receive frequency FRX.
[0078] A duplex frequency correction FDC is about equal to a difference between the nominal RF transmit frequency FTXN and the selected RF transmit frequency FTXS. As such, the duplex frequency correction FDC is about equal to a difference between the preferred notch frequency FPN and the RF duplex frequency FDP. In one embodiment of the selected RF transmit frequency
FTXS, the selected RF transmit frequency FTXS is greater than the nominal RF transmit frequency FTXN, as shown in Figure 17. However, in an alternate embodiment of the selected RF transmit frequency FTXS, the selected RF transmit frequency FTXS is less than the nominal RF transmit frequency FTXN. Therefore, the duplex frequency correction FDC may be positive or negative.
[0079] Some of the circuitry previously described may use discrete circuitry, integrated circuitry, programmable circuitry, non-volatile circuitry, volatile circuitry, software executing instructions on computing hardware, firmware executing instructions on computing hardware, the like, or any combination thereof. The computing hardware may include mainframes, micro-processors, microcontrollers, DSPs, the like, or any combination thereof.
[0080] None of the embodiments of the present disclosure are intended to limit the scope of any other embodiment of the present disclosure. Any or all of any embodiment of the present disclosure may be combined with any or all of any other embodiment of the present disclosure to create new embodiments of the present disclosure.
[0081] Those skilled in the art will recognize improvements and modifications to the embodiments of the present disclosure. All such improvements and modifications are considered within the scope of the concepts disclosed herein and the claims that follow.
Claims
1 . Circuitry comprising:
· a parallel amplifier having a parallel amplifier output and configured to regulate a voltage of a first power supply output signal based on a voltage setpoint;
• a switching supply coupled to the parallel amplifier output; and
• a radio frequency (RF) notch filter having a selectable notch frequency and coupled between the parallel amplifier output and a ground, wherein the selectable notch frequency is based on an RF duplex frequency.
2. The circuitry of claim 1 wherein the RF notch filter is a programmable RF notch filter.
3. The circuitry of claim 1 wherein the RF duplex frequency is about equal to a difference between an RF transmit frequency and an RF receive frequency.
4. The circuitry of claim 1 wherein the selectable notch frequency is about equal to the RF duplex frequency.
5. The circuitry of claim 1 wherein transmitter control circuitry is configured to select the selectable notch frequency based on the RF duplex frequency.
6. The circuitry of claim 5 further comprising the transmitter control circuitry.
7. The circuitry of claim 5 wherein selection of the selectable notch frequency is further based on notch frequency information, which is based on the RF duplex frequency.
8. The circuitry of claim 1 further comprising an offset capacitive element, wherein the switching supply is coupled to the parallel amplifier output via the offset capacitive element, and the RF notch filter is coupled to the parallel amplifier output via the offset capacitive element.
9. The circuitry of claim 1 wherein the selectable notch frequency is one of a first frequency and a second frequency.
10. The circuitry of claim 9 wherein the first frequency is equal to about 30 megahertz and the second frequency is equal to about 45 megahertz.
1 1 . The circuitry of claim 1 wherein the selectable notch frequency is one of a first frequency, a second frequency, and a third frequency.
12. The circuitry of claim 1 1 wherein the first frequency is equal to about 30 megahertz, the second frequency is equal to about 39 megahertz, and the third frequency is equal to about 47 megahertz.
13. The circuitry of claim 1 wherein the RF notch filter is configured to be disabled.
14. The circuitry of claim 1 wherein:
• the parallel amplifier is further configured to partially provide the first power supply output signal via the parallel amplifier output based on the voltage setpoint; and
• the switching supply is configured to partially provide the first power supply output signal.
15. The circuitry of claim 14 wherein the switching supply is further configured to regulate the first power supply output signal to minimize an output current from the parallel amplifier.
1 6. The circuitry of claim 1 further comprising a direct current (DC)-DC converter, an RF PA, and RF front-end circuitry, wherein:
• the DC-DC converter comprises the parallel amplifier, the switching supply, and the RF notch filter, and is configured to provide a first envelope power supply signal to the RF PA, such that the first envelope power supply signal is configured to at least partially envelope track an RF transmit signal;
• the RF PA is configured to receive and amplify an RF input signal to provide the RF transmit signal using the first envelope power supply signal; and
• the RF front-end circuitry is configured to provide an RF receive signal, which has an RF receive frequency.
17. The circuitry of claim 16 wherein the first envelope power supply signal provides power for amplification to the RF PA.
18. The circuitry of claim 16 wherein a bandwidth of the first envelope power supply signal is greater than or equal to about 20 megahertz.
19. The circuitry of claim 16 wherein the selectable notch frequency is selected to reduce noise at the RF receive frequency.
20. The circuitry of claim 1 wherein the selectable notch frequency is selected to minimize a difference between the selectable notch frequency and the RF duplex frequency.
21 . The circuitry of claim 1 wherein the selectable notch frequency is selected to minimize a difference between the selectable notch frequency and a preferred notch frequency.
22. A method comprising:
• providing a first envelope power supply signal to a radio frequency (RF) power amplifier (PA);
• at least partially envelope tracking an RF transmit signal;
• receiving and amplifying an RF input signal to provide the RF transmit signal using the first envelope power supply signal;
• providing an RF receive signal, which has an RF receive frequency; and
• RF notch filtering the first envelope power supply signal to reduce noise at the RF receive frequency.
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Families Citing this family (77)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US9099961B2 (en) | 2010-04-19 | 2015-08-04 | Rf Micro Devices, Inc. | Output impedance compensation of a pseudo-envelope follower power management system |
US9431974B2 (en) | 2010-04-19 | 2016-08-30 | Qorvo Us, Inc. | Pseudo-envelope following feedback delay compensation |
US8493141B2 (en) | 2010-04-19 | 2013-07-23 | Rf Micro Devices, Inc. | Pseudo-envelope following power management system |
WO2012047738A1 (en) | 2010-09-29 | 2012-04-12 | Rf Micro Devices, Inc. | SINGLE μC-BUCKBOOST CONVERTER WITH MULTIPLE REGULATED SUPPLY OUTPUTS |
US9246460B2 (en) | 2011-05-05 | 2016-01-26 | Rf Micro Devices, Inc. | Power management architecture for modulated and constant supply operation |
US9247496B2 (en) | 2011-05-05 | 2016-01-26 | Rf Micro Devices, Inc. | Power loop control based envelope tracking |
US9379667B2 (en) | 2011-05-05 | 2016-06-28 | Rf Micro Devices, Inc. | Multiple power supply input parallel amplifier based envelope tracking |
US9263996B2 (en) | 2011-07-20 | 2016-02-16 | Rf Micro Devices, Inc. | Quasi iso-gain supply voltage function for envelope tracking systems |
US9484797B2 (en) | 2011-10-26 | 2016-11-01 | Qorvo Us, Inc. | RF switching converter with ripple correction |
US9294041B2 (en) | 2011-10-26 | 2016-03-22 | Rf Micro Devices, Inc. | Average frequency control of switcher for envelope tracking |
US9250643B2 (en) | 2011-11-30 | 2016-02-02 | Rf Micro Devices, Inc. | Using a switching signal delay to reduce noise from a switching power supply |
US9515621B2 (en) | 2011-11-30 | 2016-12-06 | Qorvo Us, Inc. | Multimode RF amplifier system |
US9041365B2 (en) | 2011-12-01 | 2015-05-26 | Rf Micro Devices, Inc. | Multiple mode RF power converter |
US9256234B2 (en) | 2011-12-01 | 2016-02-09 | Rf Micro Devices, Inc. | Voltage offset loop for a switching controller |
US9280163B2 (en) | 2011-12-01 | 2016-03-08 | Rf Micro Devices, Inc. | Average power tracking controller |
US9494962B2 (en) | 2011-12-02 | 2016-11-15 | Rf Micro Devices, Inc. | Phase reconfigurable switching power supply |
US9813036B2 (en) | 2011-12-16 | 2017-11-07 | Qorvo Us, Inc. | Dynamic loadline power amplifier with baseband linearization |
US9298198B2 (en) | 2011-12-28 | 2016-03-29 | Rf Micro Devices, Inc. | Noise reduction for envelope tracking |
US9225231B2 (en) | 2012-09-14 | 2015-12-29 | Rf Micro Devices, Inc. | Open loop ripple cancellation circuit in a DC-DC converter |
US9207692B2 (en) | 2012-10-18 | 2015-12-08 | Rf Micro Devices, Inc. | Transitioning from envelope tracking to average power tracking |
US9627975B2 (en) | 2012-11-16 | 2017-04-18 | Qorvo Us, Inc. | Modulated power supply system and method with automatic transition between buck and boost modes |
WO2014116933A2 (en) | 2013-01-24 | 2014-07-31 | Rf Micro Devices, Inc | Communications based adjustments of an envelope tracking power supply |
US9479118B2 (en) | 2013-04-16 | 2016-10-25 | Rf Micro Devices, Inc. | Dual instantaneous envelope tracking |
US9236957B2 (en) * | 2013-05-07 | 2016-01-12 | Rf Micro Devices, Inc. | Technique to reduce the third harmonic of an on-state RF switch |
US9374005B2 (en) | 2013-08-13 | 2016-06-21 | Rf Micro Devices, Inc. | Expanded range DC-DC converter |
US9614476B2 (en) | 2014-07-01 | 2017-04-04 | Qorvo Us, Inc. | Group delay calibration of RF envelope tracking |
GB2535178A (en) * | 2015-02-11 | 2016-08-17 | Snaptrack Inc | Switcher noise reduction |
GB2535180A (en) * | 2015-02-11 | 2016-08-17 | Snaptrack Inc | AC amplifier output impedance reduction |
WO2016138379A1 (en) * | 2015-02-27 | 2016-09-01 | University Of Georgia Research Foundation, Inc. | Ultra high-speed photonics based radio frequency switching |
US9941844B2 (en) | 2015-07-01 | 2018-04-10 | Qorvo Us, Inc. | Dual-mode envelope tracking power converter circuitry |
US9912297B2 (en) | 2015-07-01 | 2018-03-06 | Qorvo Us, Inc. | Envelope tracking power converter circuitry |
CN105071055B (en) * | 2015-07-28 | 2018-12-28 | 福建联迪商用设备有限公司 | A kind of the RF antenna impedance matching and its design method of anti-High-frequency Interference |
US9712197B2 (en) | 2015-08-28 | 2017-07-18 | Skyworks Solutions, Inc. | Tunable notch filter and contour tuning circuit |
WO2017040221A1 (en) * | 2015-08-28 | 2017-03-09 | Skyworks Solutions, Inc | Tunable notch filter |
WO2017040222A1 (en) | 2015-09-02 | 2017-03-09 | Skyworks Solutions, Inc. | Contour tuning circuit |
US10373794B2 (en) | 2015-10-29 | 2019-08-06 | Lam Research Corporation | Systems and methods for filtering radio frequencies from a signal of a thermocouple and controlling a temperature of an electrode in a plasma chamber |
US10043636B2 (en) | 2015-12-10 | 2018-08-07 | Lam Research Corporation | Apparatuses and methods for avoiding electrical breakdown from RF terminal to adjacent non-RF terminal |
US10382071B2 (en) * | 2016-01-27 | 2019-08-13 | Qorvo Us, Inc. | Bandwidth optimization for power amplifier power supplies |
US9973147B2 (en) | 2016-05-10 | 2018-05-15 | Qorvo Us, Inc. | Envelope tracking power management circuit |
US10128798B2 (en) * | 2016-07-29 | 2018-11-13 | Qualcomm Incorporated | Adjusting envelope tracking power supply |
US10187122B2 (en) | 2017-02-22 | 2019-01-22 | Samsung Electronics Co., Ltd. | Near field communications device |
US10158330B1 (en) | 2017-07-17 | 2018-12-18 | Qorvo Us, Inc. | Multi-mode envelope tracking amplifier circuit |
US10680559B2 (en) | 2017-10-06 | 2020-06-09 | Qorvo Us, Inc. | Envelope tracking system for transmitting a wide modulation bandwidth signal(s) |
US10230340B1 (en) * | 2018-02-06 | 2019-03-12 | Qorvo Us, Inc. | Wide modulation bandwidth radio frequency circuit |
US10476437B2 (en) | 2018-03-15 | 2019-11-12 | Qorvo Us, Inc. | Multimode voltage tracker circuit |
US10944365B2 (en) | 2018-06-28 | 2021-03-09 | Qorvo Us, Inc. | Envelope tracking amplifier circuit |
US11088618B2 (en) | 2018-09-05 | 2021-08-10 | Qorvo Us, Inc. | PWM DC-DC converter with linear voltage regulator for DC assist |
US10911001B2 (en) | 2018-10-02 | 2021-02-02 | Qorvo Us, Inc. | Envelope tracking amplifier circuit |
US11018638B2 (en) | 2018-10-31 | 2021-05-25 | Qorvo Us, Inc. | Multimode envelope tracking circuit and related apparatus |
US10985702B2 (en) | 2018-10-31 | 2021-04-20 | Qorvo Us, Inc. | Envelope tracking system |
US10938351B2 (en) | 2018-10-31 | 2021-03-02 | Qorvo Us, Inc. | Envelope tracking system |
US10680556B2 (en) | 2018-11-05 | 2020-06-09 | Qorvo Us, Inc. | Radio frequency front-end circuit |
US11031909B2 (en) | 2018-12-04 | 2021-06-08 | Qorvo Us, Inc. | Group delay optimization circuit and related apparatus |
US11082007B2 (en) | 2018-12-19 | 2021-08-03 | Qorvo Us, Inc. | Envelope tracking integrated circuit and related apparatus |
US11146213B2 (en) | 2019-01-15 | 2021-10-12 | Qorvo Us, Inc. | Multi-radio access technology envelope tracking amplifier apparatus |
US11025458B2 (en) | 2019-02-07 | 2021-06-01 | Qorvo Us, Inc. | Adaptive frequency equalizer for wide modulation bandwidth envelope tracking |
US10998859B2 (en) | 2019-02-07 | 2021-05-04 | Qorvo Us, Inc. | Dual-input envelope tracking integrated circuit and related apparatus |
US11233481B2 (en) | 2019-02-18 | 2022-01-25 | Qorvo Us, Inc. | Modulated power apparatus |
US11374482B2 (en) | 2019-04-02 | 2022-06-28 | Qorvo Us, Inc. | Dual-modulation power management circuit |
US11082009B2 (en) | 2019-04-12 | 2021-08-03 | Qorvo Us, Inc. | Envelope tracking power amplifier apparatus |
US11018627B2 (en) | 2019-04-17 | 2021-05-25 | Qorvo Us, Inc. | Multi-bandwidth envelope tracking integrated circuit and related apparatus |
US11424719B2 (en) | 2019-04-18 | 2022-08-23 | Qorvo Us, Inc. | Multi-bandwidth envelope tracking integrated circuit |
US11031911B2 (en) | 2019-05-02 | 2021-06-08 | Qorvo Us, Inc. | Envelope tracking integrated circuit and related apparatus |
US11349436B2 (en) | 2019-05-30 | 2022-05-31 | Qorvo Us, Inc. | Envelope tracking integrated circuit |
US11632089B2 (en) * | 2019-06-20 | 2023-04-18 | Mediatek Inc. | Notch circuit and power amplifier module |
US11539289B2 (en) | 2019-08-02 | 2022-12-27 | Qorvo Us, Inc. | Multi-level charge pump circuit |
US11309922B2 (en) | 2019-12-13 | 2022-04-19 | Qorvo Us, Inc. | Multi-mode power management integrated circuit in a small formfactor wireless apparatus |
US11349513B2 (en) | 2019-12-20 | 2022-05-31 | Qorvo Us, Inc. | Envelope tracking system |
US11539330B2 (en) | 2020-01-17 | 2022-12-27 | Qorvo Us, Inc. | Envelope tracking integrated circuit supporting multiple types of power amplifiers |
US11716057B2 (en) | 2020-01-28 | 2023-08-01 | Qorvo Us, Inc. | Envelope tracking circuitry |
US11728774B2 (en) | 2020-02-26 | 2023-08-15 | Qorvo Us, Inc. | Average power tracking power management integrated circuit |
US11196392B2 (en) | 2020-03-30 | 2021-12-07 | Qorvo Us, Inc. | Device and device protection system |
US11588449B2 (en) | 2020-09-25 | 2023-02-21 | Qorvo Us, Inc. | Envelope tracking power amplifier apparatus |
US11728796B2 (en) | 2020-10-14 | 2023-08-15 | Qorvo Us, Inc. | Inverted group delay circuit |
US11909385B2 (en) | 2020-10-19 | 2024-02-20 | Qorvo Us, Inc. | Fast-switching power management circuit and related apparatus |
US20230387860A1 (en) * | 2022-05-31 | 2023-11-30 | Qorvo Us, Inc. | Voltage ripple reduction in a power management circuit |
CN115459760B (en) * | 2022-11-09 | 2023-03-03 | 青岛泰戈菲斯海洋装备股份公司 | Frequency discrimination circuit of acoustic releaser |
Citations (6)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20060178119A1 (en) * | 2005-02-09 | 2006-08-10 | Nokia Corporation | Variable bandwidth envelope modulator for use with envelope elimination and restoration transmitter architecture and method |
WO2007149346A2 (en) * | 2006-06-16 | 2007-12-27 | Pulsewave Rf, Inc. | Radio frequency power amplifier and method using a controlled supply |
US20100266066A1 (en) * | 2007-11-05 | 2010-10-21 | Nec Corporation | Power amplifier and radio wave transmitter having the same |
US20120025907A1 (en) * | 2010-07-28 | 2012-02-02 | Korea Advanced Institute Of Science And Technology | Power amplifier |
GB2484475A (en) * | 2010-10-11 | 2012-04-18 | Toshiba Res Europ Ltd | A power supply modulator for an RF amplifier, using a current-output class G amplifier |
US20120154054A1 (en) * | 2010-12-17 | 2012-06-21 | Skyworks Solutions, Inc. | Apparatus and methods for oscillation suppression |
Family Cites Families (271)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3980964A (en) | 1974-05-20 | 1976-09-14 | Grodinsky Robert M | Noise reduction circuit |
US3969682A (en) | 1974-10-21 | 1976-07-13 | Oberheim Electronics Inc. | Circuit for dynamic control of phase shift |
US4587552A (en) | 1983-09-02 | 1986-05-06 | Rca Corporation | Apparatus for generating the magnitude of the vector sum of two orthogonal signals as for use in a digital TV receiver |
US4692889A (en) | 1984-09-28 | 1987-09-08 | Rca Corporation | Circuitry for calculating magnitude of vector sum from its orthogonal components in digital television receiver |
US4831258A (en) | 1988-03-04 | 1989-05-16 | Exergen Corporation | Dual sensor radiation detector |
US4996500A (en) | 1989-10-24 | 1991-02-26 | Hewlett-Packard Company | Automatic control system |
US5311309A (en) | 1990-06-01 | 1994-05-10 | Thomson Consumer Electronics, Inc. | Luminance processing system for compressing and expanding video data |
US5486871A (en) | 1990-06-01 | 1996-01-23 | Thomson Consumer Electronics, Inc. | Automatic letterbox detection |
US5420643A (en) | 1990-06-01 | 1995-05-30 | Thomson Consumer Electronics, Inc. | Chrominance processing system for compressing and expanding video data |
US5351087A (en) | 1990-06-01 | 1994-09-27 | Thomson Consumer Electronics, Inc. | Two stage interpolation system |
US5099203A (en) | 1990-06-05 | 1992-03-24 | Continental Electronics Corporation | Power amplifier having multiple switched stages and method of operating same |
DE4038111A1 (en) | 1990-11-29 | 1992-06-04 | Thomson Brandt Gmbh | UNIVERSAL FILTER |
US5146504A (en) | 1990-12-07 | 1992-09-08 | Motorola, Inc. | Speech selective automatic gain control |
US5187396A (en) | 1991-05-22 | 1993-02-16 | Benchmarq Microelectronics, Inc. | Differential comparator powered from signal input terminals for use in power switching applications |
JPH0828965B2 (en) | 1992-09-02 | 1996-03-21 | 日本電気株式会社 | Voltage conversion circuit |
US5457620A (en) | 1993-07-30 | 1995-10-10 | At&T Ipm Corp. | Current estimating circuit for switch mode power supply |
US5414614A (en) | 1994-06-06 | 1995-05-09 | Motorola, Inc. | Dynamically configurable switched capacitor power supply and method |
US5822318A (en) | 1994-07-29 | 1998-10-13 | Qualcomm Incorporated | Method and apparatus for controlling power in a variable rate communication system |
US5646621A (en) | 1994-11-02 | 1997-07-08 | Advanced Micro Devices, Inc. | Delta-sigma ADC with multi-stage decimation filter and gain compensation filter |
US5581454A (en) | 1994-11-22 | 1996-12-03 | Collins; Hansel | High power switched capacitor voltage conversion and regulation apparatus |
US5541547A (en) | 1995-05-03 | 1996-07-30 | Sun Microsystems, Inc. | Test generator system for controllably inducing power pin latch-up and signal pin latch-up in a CMOS device |
JP3110288B2 (en) | 1995-07-21 | 2000-11-20 | 日本電気株式会社 | Exponential logarithmic conversion circuit |
US5715526A (en) | 1995-09-08 | 1998-02-03 | Qualcomm Incorporated | Apparatus and method for controlling transmission power in a cellular communications system |
US5767744A (en) | 1995-11-22 | 1998-06-16 | Qsc Audio Products, Inc. | Lightweight fixed frequency discontinuous resonant power supply for audio amplifiers |
US5732333A (en) | 1996-02-14 | 1998-03-24 | Glenayre Electronics, Inc. | Linear transmitter using predistortion |
US6256482B1 (en) | 1997-04-07 | 2001-07-03 | Frederick H. Raab | Power- conserving drive-modulation method for envelope-elimination-and-restoration (EER) transmitters |
US5905407A (en) | 1997-07-30 | 1999-05-18 | Motorola, Inc. | High efficiency power amplifier using combined linear and switching techniques with novel feedback system |
US5936464A (en) | 1997-11-03 | 1999-08-10 | Motorola, Inc. | Method and apparatus for reducing distortion in a high efficiency power amplifier |
US6141541A (en) | 1997-12-31 | 2000-10-31 | Motorola, Inc. | Method, device, phone and base station for providing envelope-following for variable envelope radio frequency signals |
FR2773423B1 (en) | 1998-01-06 | 2001-10-19 | Alsthom Cge Alkatel | METHOD AND SYSTEM FOR DIGITAL LINEARIZATION OF AN AMPLIFIER |
US5898342A (en) | 1998-01-20 | 1999-04-27 | Advanced Micro Devices | Power amplifier arrangement and method for data signal interface |
US6055168A (en) | 1998-03-04 | 2000-04-25 | National Semiconductor Corporation | Capacitor DC-DC converter with PFM and gain hopping |
FR2776144B1 (en) | 1998-03-13 | 2000-07-13 | Sgs Thomson Microelectronics | CIRCUIT FOR SWITCHING ANALOG SIGNALS OF AMPLITUDES HIGHER THAN THE SUPPLY VOLTAGE |
US6070181A (en) | 1998-03-27 | 2000-05-30 | Chun-Shan Institute Of Science And Technology | Method and circuit for envelope detection using a peel cone approximation |
US6198645B1 (en) | 1998-07-02 | 2001-03-06 | National Semiconductor Corporation | Buck and boost switched capacitor gain stage with optional shared rest state |
US6043610A (en) | 1998-07-16 | 2000-03-28 | Durel Corporation | Battery operated power supply including a low level boost and a high level boost |
US6690652B1 (en) | 1998-10-26 | 2004-02-10 | International Business Machines Corporation | Adaptive power control in wideband CDMA cellular systems (WCDMA) and methods of operation |
JP3144398B2 (en) | 1998-10-27 | 2001-03-12 | 日本電気株式会社 | Variable delay circuit |
DE69930453T2 (en) * | 1998-10-27 | 2006-09-28 | Murata Manufacturing Co., Ltd., Nagaokakyo | Composite high frequency component and mobile communication device equipped therewith |
SG90712A1 (en) | 1998-12-05 | 2002-08-20 | Inst Of Microelectronics | Power amplifier |
US6043707A (en) | 1999-01-07 | 2000-03-28 | Motorola, Inc. | Method and apparatus for operating a radio-frequency power amplifier as a variable-class linear amplifier |
US6864668B1 (en) | 1999-02-09 | 2005-03-08 | Tropian, Inc. | High-efficiency amplifier output level and burst control |
US6377784B2 (en) | 1999-02-09 | 2002-04-23 | Tropian, Inc. | High-efficiency modulation RF amplifier |
US6118343A (en) | 1999-05-10 | 2000-09-12 | Tyco Electronics Logistics Ag | Power Amplifier incorporating single drain switch and single negative voltage generator |
US6701141B2 (en) | 1999-05-18 | 2004-03-02 | Lockheed Martin Corporation | Mixed signal true time delay digital beamformer |
US6621808B1 (en) | 1999-08-13 | 2003-09-16 | International Business Machines Corporation | Adaptive power control based on a rake receiver configuration in wideband CDMA cellular systems (WCDMA) and methods of operation |
FR2798014B1 (en) | 1999-08-31 | 2002-03-29 | St Microelectronics Sa | SUPPLY CIRCUIT WITH VOLTAGE SELECTOR |
US6147478A (en) | 1999-09-17 | 2000-11-14 | Texas Instruments Incorporated | Hysteretic regulator and control method having switching frequency independent from output filter |
US6681101B1 (en) | 2000-01-11 | 2004-01-20 | Skyworks Solutions, Inc. | RF transmitter with extended efficient power control range |
US6452366B1 (en) * | 2000-02-11 | 2002-09-17 | Champion Microelectronic Corp. | Low power mode and feedback arrangement for a switching power converter |
US6300826B1 (en) | 2000-05-05 | 2001-10-09 | Ericsson Telefon Ab L M | Apparatus and method for efficiently amplifying wideband envelope signals |
TW480415B (en) | 2000-05-17 | 2002-03-21 | Chung Shan Inst Of Science | Demodulation apparatus of square root and method of the same |
US6654594B1 (en) | 2000-05-30 | 2003-11-25 | Motorola, Inc. | Digitized automatic gain control system and methods for a controlled gain receiver |
JP2002076951A (en) | 2000-08-25 | 2002-03-15 | Sharp Corp | Power supply circuit for transmitter |
US6348780B1 (en) | 2000-09-22 | 2002-02-19 | Texas Instruments Incorporated | Frequency control of hysteretic power converter by adjusting hystersis levels |
CN1157880C (en) | 2000-09-25 | 2004-07-14 | 华为技术有限公司 | Multiple time interval power control method |
US6559689B1 (en) | 2000-10-02 | 2003-05-06 | Allegro Microsystems, Inc. | Circuit providing a control voltage to a switch and including a capacitor |
US6975686B1 (en) | 2000-10-31 | 2005-12-13 | Telefonaktiebolaget L.M. Ericsson | IQ modulation systems and methods that use separate phase and amplitude signal paths |
US6583610B2 (en) | 2001-03-12 | 2003-06-24 | Semtech Corporation | Virtual ripple generation in switch-mode power supplies |
US7010276B2 (en) | 2001-04-11 | 2006-03-07 | Tropian, Inc. | Communications signal amplifiers having independent power control and amplitude modulation |
US6819938B2 (en) | 2001-06-26 | 2004-11-16 | Qualcomm Incorporated | System and method for power control calibration and a wireless communication device |
US6707865B2 (en) | 2001-07-16 | 2004-03-16 | Qualcomm Incorporated | Digital voltage gain amplifier for zero IF architecture |
US6731694B2 (en) | 2001-08-07 | 2004-05-04 | Motorola, Inc. | Isolator eliminator for a linear transmitter |
US6781452B2 (en) | 2001-08-29 | 2004-08-24 | Tropian, Inc. | Power supply processing for power amplifiers |
US7164893B2 (en) | 2001-08-31 | 2007-01-16 | Motorola, Inc. | Method and apparatus for optimizing supply modulation in a transmitter |
JP2003124821A (en) | 2001-09-28 | 2003-04-25 | Motorola Inc | Transmitting power control circuit |
US7031457B2 (en) | 2001-11-30 | 2006-04-18 | Texas Instruments Incorporated | Programmable peak detector for use with zero-overhead Class G line drivers |
JP3932259B2 (en) * | 2001-12-12 | 2007-06-20 | 株式会社ルネサステクノロジ | High frequency power amplifier circuit and electronic component for wireless communication |
US6661210B2 (en) | 2002-01-23 | 2003-12-09 | Telfonaktiebolaget L.M. Ericsson | Apparatus and method for DC-to-DC power conversion |
US6788151B2 (en) | 2002-02-06 | 2004-09-07 | Lucent Technologies Inc. | Variable output power supply |
KR100832117B1 (en) | 2002-02-17 | 2008-05-27 | 삼성전자주식회사 | Apparatus for transmitting/receiving uplink power offset in communication system using high speed downlink packet access scheme |
US7254157B1 (en) | 2002-03-27 | 2007-08-07 | Xilinx, Inc. | Method and apparatus for generating a phase locked spread spectrum clock signal |
US6643148B1 (en) * | 2002-04-18 | 2003-11-04 | Alcatel Canada Inc. | Audio band conducted emissions suppression on power feeders |
US7158586B2 (en) | 2002-05-03 | 2007-01-02 | Atheros Communications, Inc. | Systems and methods to provide wideband magnitude and phase imbalance calibration and compensation in quadrature receivers |
US7171435B2 (en) | 2002-05-17 | 2007-01-30 | Texas Instruments Incorporated | Circuits, systems, and methods implementing approximations for logarithm, inverse logarithm, and reciprocal |
US6703080B2 (en) | 2002-05-20 | 2004-03-09 | Eni Technology, Inc. | Method and apparatus for VHF plasma processing with load mismatch reliability and stability |
US7233624B2 (en) | 2002-06-11 | 2007-06-19 | Interdigital Technology Corporation | Method and system for all digital gain control |
US6624712B1 (en) | 2002-06-11 | 2003-09-23 | Motorola, Inc. | Method and apparatus for power modulating to prevent instances of clipping |
US6725021B1 (en) | 2002-06-20 | 2004-04-20 | Motorola, Inc. | Method for tuning an envelope tracking amplification system |
ITTO20020545A1 (en) | 2002-06-21 | 2003-12-22 | St Microelectronics Srl | CONTROL CIRCUIT IN PWM MODE FOR THE POST-REGULATION OF SWITCHING POWER SUPPLIES WITH MANY OUTPUTS |
JP2004064937A (en) | 2002-07-31 | 2004-02-26 | Nec Corp | Charge pump-type boosting circuit |
US6728163B2 (en) | 2002-08-23 | 2004-04-27 | Micron Technology, Inc. | Controlling a delay lock loop circuit |
US6744151B2 (en) | 2002-09-13 | 2004-06-01 | Analog Devices, Inc. | Multi-channel power supply selector |
US7263135B2 (en) | 2002-10-03 | 2007-08-28 | Matsushita Electric Industrial Co., Ltd. | Transmitting method and transmitter apparatus |
KR20050053591A (en) | 2002-10-28 | 2005-06-08 | 마츠시타 덴끼 산교 가부시키가이샤 | Transmitter |
US6958596B1 (en) * | 2002-12-20 | 2005-10-25 | Intersil Americas Inc. | Compensation sample and hold for voltage regulator amplifier |
US6801082B2 (en) | 2002-12-31 | 2004-10-05 | Motorola, Inc. | Power amplifier circuit and method using bandlimited signal component estimates |
US7206557B2 (en) | 2003-01-08 | 2007-04-17 | Lucent Technologies Inc. | Method and apparatus for suppressing local oscillator leakage in a wireless transmitter |
GB2398648B (en) | 2003-02-19 | 2005-11-09 | Nujira Ltd | Power supply stage for an amplifier |
ATE551773T1 (en) | 2003-02-20 | 2012-04-15 | Sony Ericsson Mobile Comm Ab | EFFICIENT MODULATION OF HIGH FREQUENCY SIGNALS |
US7193470B2 (en) | 2003-03-04 | 2007-03-20 | Samsung Electronics Co., Ltd. | Method and apparatus for controlling a power amplifier in a mobile communication system |
EP2284996A1 (en) | 2003-03-12 | 2011-02-16 | MediaTek Inc. | Closed loop power control of non-constant envelope waveforms using sample/hold |
US7907010B2 (en) | 2003-04-07 | 2011-03-15 | Nxp B.V. | Digital amplifier |
JP3972856B2 (en) * | 2003-04-16 | 2007-09-05 | 富士電機ホールディングス株式会社 | Power system |
US7072626B2 (en) | 2003-04-30 | 2006-07-04 | Telefonaktiebolaget Lm Ericsson (Publ) | Polar modulation transmitter |
US7805115B1 (en) | 2003-06-02 | 2010-09-28 | Analog Devices, Inc. | Variable filter systems and methods for enhanced data rate communication systems |
US7321912B2 (en) | 2003-06-24 | 2008-01-22 | Texas Instruments Incorporated | Device with dB-to-linear gain conversion |
US7043213B2 (en) | 2003-06-24 | 2006-05-09 | Northrop Grumman Corporation | Multi-mode amplifier system |
US7251462B2 (en) | 2003-07-08 | 2007-07-31 | Matsushita Electric Industrial Co., Ltd. | Modulation circuit device, modulation method and radio communication device |
FR2857532B1 (en) | 2003-07-08 | 2005-08-19 | Thales Sa | METHOD OF ESTIMATING CARRIER RESIDUE, ESTIMATOR AND MODULATION SYSTEM WITH CARRIER LOADING USING THE SAME |
US7043518B2 (en) | 2003-07-31 | 2006-05-09 | Cradle Technologies, Inc. | Method and system for performing parallel integer multiply accumulate operations on packed data |
KR100602065B1 (en) | 2003-07-31 | 2006-07-14 | 엘지전자 주식회사 | Power supply and driving method thereof and driving apparatus and method using the electro-luminescence display device |
US7170341B2 (en) | 2003-08-05 | 2007-01-30 | Motorola, Inc. | Low power consumption adaptive power amplifier |
US20050032499A1 (en) | 2003-08-08 | 2005-02-10 | Cho Jin Wook | Radio frequency power detecting circuit and method therefor |
KR100524985B1 (en) | 2003-08-26 | 2005-10-31 | 삼성전자주식회사 | Effective boosting circuit, boosting power unit having it and providing for automatically load-dependent boosting, and power boosting control method thereof |
JP4589665B2 (en) * | 2003-08-29 | 2010-12-01 | ルネサスエレクトロニクス株式会社 | Amplifier and high-frequency power amplifier using the same |
US7058373B2 (en) | 2003-09-16 | 2006-06-06 | Nokia Corporation | Hybrid switched mode/linear power amplifier power supply for use in polar transmitter |
US7053718B2 (en) | 2003-09-25 | 2006-05-30 | Silicon Laboratories Inc. | Stacked RF power amplifier |
US6903608B2 (en) | 2003-10-30 | 2005-06-07 | Sige Semiconductor Inc. | Power level controlling of first amplification stage for an integrated RF power amplifier |
US7627622B2 (en) | 2003-11-14 | 2009-12-01 | International Business Machines Corporation | System and method of curve fitting |
US7026868B2 (en) | 2003-11-20 | 2006-04-11 | Northrop Grumman Corporation | Variable supply amplifier system |
US6995995B2 (en) | 2003-12-03 | 2006-02-07 | Fairchild Semiconductor Corporation | Digital loop for regulating DC/DC converter with segmented switching |
JP2005175561A (en) | 2003-12-08 | 2005-06-30 | Renesas Technology Corp | Power supply circuit for high frequency power amplifier circuit, semiconductor integrated circuit for power supply, and electronic component for power supply |
US7330501B2 (en) | 2004-01-15 | 2008-02-12 | Broadcom Corporation | Orthogonal normalization for a radio frequency integrated circuit |
US7915954B2 (en) | 2004-01-16 | 2011-03-29 | Qualcomm, Incorporated | Amplifier predistortion and autocalibration method and apparatus |
US6958594B2 (en) * | 2004-01-21 | 2005-10-25 | Analog Devices, Inc. | Switched noise filter circuit for a DC-DC converter |
WO2005076467A1 (en) * | 2004-02-06 | 2005-08-18 | Mitsubishi Denki Kabushiki Kaisha | Power amplifier unit, communication terminal and control method of power amplifier unit |
US7595569B2 (en) | 2004-02-17 | 2009-09-29 | Agere Systems Inc. | Versatile and intelligent power controller |
ES2265122T3 (en) | 2004-02-20 | 2007-02-01 | Research In Motion Limited | METHOD AND APPLIANCE TO IMPROVE THE EFFICIENCY OF POWER AMPLIFICATION IN WIRELESS COMMUNICATION SYSTEMS WITH HIGH RELATIONSHIPS BETWEEN THE VALUES OF PEAK POWER AND AVERAGE POWER. |
US7408979B2 (en) | 2004-06-28 | 2008-08-05 | Broadcom Corporation | Integrated radio circuit having multiple function I/O modules |
US7358806B2 (en) * | 2004-07-08 | 2008-04-15 | Amalfi Semiconductor, Inc. | Method and apparatus for an improved power amplifier |
US7253589B1 (en) | 2004-07-09 | 2007-08-07 | National Semiconductor Corporation | Dual-source CMOS battery charger |
US7529523B1 (en) | 2004-08-23 | 2009-05-05 | Rf Micro Devices, Inc. | N-th order curve fit for power calibration in a mobile terminal |
GB0418944D0 (en) | 2004-08-25 | 2004-09-29 | Siemens Ag | Method for envelope clipping |
JP4574471B2 (en) | 2004-09-17 | 2010-11-04 | 株式会社日立国際電気 | Distortion compensated quadrature modulator and radio transmitter |
US7378828B2 (en) * | 2004-11-09 | 2008-05-27 | The Boeing Company | DC-DC converter having magnetic feedback |
US7394233B1 (en) * | 2004-12-02 | 2008-07-01 | Nortel Networks Limited | High efficiency modulated power supply |
US7539466B2 (en) | 2004-12-14 | 2009-05-26 | Motorola, Inc. | Amplifier with varying supply voltage and input attenuation based upon supply voltage |
EP1829229B1 (en) * | 2004-12-22 | 2019-01-23 | Nokia Technologies Oy | Interoperability improvement between receivers and transmitters in a mobile station |
EP1834460A1 (en) | 2004-12-27 | 2007-09-19 | Koninklijke Philips Electronics N.V. | Transmitter apparatus |
DE602006010972D1 (en) | 2005-01-06 | 2010-01-21 | Panasonic Corp | Polar modulator and wireless communication device with it |
TWI281305B (en) | 2005-02-03 | 2007-05-11 | Richtek Techohnology Corp | Dual input voltage converter and its control method |
US20060181340A1 (en) | 2005-02-17 | 2006-08-17 | Zywyn Corporation | Regulating charge pump |
US20060199553A1 (en) | 2005-03-07 | 2006-09-07 | Andrew Corporation | Integrated transceiver with envelope tracking |
KR100588334B1 (en) * | 2005-03-29 | 2006-06-09 | 삼성전자주식회사 | Dc-dc converter using pseudo schmitt trigger circuit and method of pulse width modulation |
JP2008537467A (en) * | 2005-04-20 | 2008-09-11 | エヌエックスピー ビー ヴィ | Parallel arranged linear amplifier and DC-DC converter |
US7773691B2 (en) | 2005-04-25 | 2010-08-10 | Rf Micro Devices, Inc. | Power control system for a continuous time mobile transmitter |
TWI293828B (en) | 2005-04-28 | 2008-02-21 | Novatek Microelectronics Corp | Charge pump |
US7348847B2 (en) | 2005-04-28 | 2008-03-25 | Sige Semiconductor Inc. | Integrated implementation of a collector boost scheme and method therefor |
US7279875B2 (en) * | 2005-06-16 | 2007-10-09 | Ge Gan | High switching frequency DC-DC converter with fast response time |
DE102005030123B4 (en) | 2005-06-28 | 2017-08-31 | Austriamicrosystems Ag | Power supply arrangement and its use |
US7283406B2 (en) | 2005-07-11 | 2007-10-16 | Taiwan Semiconductor Manufacturing Co., Ltd. | High voltage wordline driver with a three stage level shifter |
US20070014382A1 (en) | 2005-07-15 | 2007-01-18 | Nokia Corporation | Reconfigurable transmitter |
KR20080031463A (en) | 2005-07-27 | 2008-04-08 | 엔엑스피 비 브이 | Rf transmitter with compensation of differential path delay |
US7602155B2 (en) | 2005-07-27 | 2009-10-13 | Artesyn Technologies, Inc. | Power supply providing ultrafast modulation of output voltage |
US7233130B1 (en) | 2005-08-05 | 2007-06-19 | Rf Micro Devices, Inc. | Active ripple reduction switched mode power supplies |
US20070063681A1 (en) | 2005-09-16 | 2007-03-22 | Amazion Electronics, Inc. | Direct mode pulse width modulation for DC to DC converters |
US7615979B2 (en) | 2005-11-28 | 2009-11-10 | David J. Caldwell | Flexible power converter and simplified process controller |
JP5003134B2 (en) | 2006-01-10 | 2012-08-15 | 日本電気株式会社 | Amplifier |
US7512395B2 (en) | 2006-01-31 | 2009-03-31 | International Business Machines Corporation | Receiver and integrated AM-FM/IQ demodulators for gigabit-rate data detection |
JP2007209103A (en) * | 2006-02-01 | 2007-08-16 | Ricoh Co Ltd | Current mode control dc-dc converter |
US7522676B2 (en) | 2006-02-06 | 2009-04-21 | Nokia Corporation | Method and system for transmitter envelope delay calibration |
TWI309102B (en) | 2006-03-02 | 2009-04-21 | Himax Tech Inc | A voltage switch apparatus |
US8648579B2 (en) | 2006-03-17 | 2014-02-11 | St-Ericsson Sa | Supply circuit with ripple compensation |
US7826810B2 (en) | 2006-05-08 | 2010-11-02 | Harris Corporation | Multiband radio with transmitter output power optimization |
US7873331B2 (en) | 2006-06-04 | 2011-01-18 | Samsung Electro-Mechanics Company, Ltd. | Systems, methods, and apparatuses for multi-path orthogonal recursive predistortion |
JP2008035487A (en) | 2006-06-19 | 2008-02-14 | Renesas Technology Corp | Rf power amplifier |
GB2446843B (en) | 2006-06-30 | 2011-09-07 | Wolfson Microelectronics Plc | Amplifier circuit and methods of operation thereof |
FI20065457A0 (en) | 2006-06-30 | 2006-06-30 | Nokia Corp | Power amplifier switching power supply control |
US8311243B2 (en) | 2006-08-21 | 2012-11-13 | Cirrus Logic, Inc. | Energy-efficient consumer device audio power output stage |
US8068622B2 (en) | 2006-12-13 | 2011-11-29 | Cirrus Logic, Inc. | Method and apparatus for controlling a selectable voltage audio power output stage |
US7729670B2 (en) | 2006-09-29 | 2010-06-01 | Broadcom Corporation | Method and system for minimizing power consumption in a communication system |
US7646108B2 (en) | 2006-09-29 | 2010-01-12 | Intel Corporation | Multiple output voltage regulator |
US7454238B2 (en) | 2006-10-30 | 2008-11-18 | Quantance, Inc. | Power combining power supply system |
US7856048B1 (en) | 2006-11-20 | 2010-12-21 | Marvell International, Ltd. | On-chip IQ imbalance and LO leakage calibration for transceivers |
KR100794310B1 (en) | 2006-11-21 | 2008-01-11 | 삼성전자주식회사 | Switched capacitor circuit and amplifing method thereof |
US7777470B2 (en) | 2006-12-06 | 2010-08-17 | Texas Instruments Incorporated | System and method for controlling a hysteretic mode converter |
WO2008072134A1 (en) | 2006-12-12 | 2008-06-19 | Koninklijke Philips Electronics N.V. | A high efficiency modulating rf amplifier |
US7986931B2 (en) | 2006-12-12 | 2011-07-26 | Industrial Technology Research Institute | RFID reader and circuit and method for echo cancellation thereof |
GB2444984B (en) | 2006-12-22 | 2011-07-13 | Wolfson Microelectronics Plc | Charge pump circuit and methods of operation thereof |
US7777459B2 (en) | 2006-12-30 | 2010-08-17 | Advanced Analogic Technologies, Inc. | High-efficiency DC/DC voltage converter including capacitive switching pre-converter and down inductive switching post-regulator |
US7675365B2 (en) | 2007-01-10 | 2010-03-09 | Samsung Electro-Mechanics | Systems and methods for power amplifiers with voltage boosting multi-primary transformers |
WO2008090721A1 (en) | 2007-01-24 | 2008-07-31 | Nec Corporation | Power amplifier |
US7679433B1 (en) | 2007-02-02 | 2010-03-16 | National Semiconductor Corporation | Circuit and method for RF power amplifier power regulation and modulation envelope tracking |
EP1962413A1 (en) | 2007-02-22 | 2008-08-27 | Stmicroelectronics SA | Ripple compensator and switching converter comprising such a ripple compensator |
US7859336B2 (en) | 2007-03-13 | 2010-12-28 | Astec International Limited | Power supply providing ultrafast modulation of output voltage |
KR101309293B1 (en) | 2007-03-28 | 2013-09-16 | 페어차일드코리아반도체 주식회사 | Switching mode power supply and the driving method thereof |
US7696735B2 (en) | 2007-03-30 | 2010-04-13 | Intel Corporation | Switched capacitor converters |
US7791324B2 (en) | 2007-03-30 | 2010-09-07 | Intersil Americas Inc. | Switching regulator without a dedicated input current sense element |
US8274332B2 (en) | 2007-04-23 | 2012-09-25 | Dali Systems Co. Ltd. | N-way Doherty distributed power amplifier with power tracking |
US7554473B2 (en) | 2007-05-02 | 2009-06-30 | Cirrus Logic, Inc. | Control system using a nonlinear delta-sigma modulator with nonlinear process modeling |
US7466195B2 (en) | 2007-05-18 | 2008-12-16 | Quantance, Inc. | Error driven RF power amplifier control with increased efficiency |
US20090004981A1 (en) | 2007-06-27 | 2009-01-01 | Texas Instruments Incorporated | High efficiency digital transmitter incorporating switching power supply and linear power amplifier |
GB0715254D0 (en) | 2007-08-03 | 2007-09-12 | Wolfson Ltd | Amplifier circuit |
US7609114B2 (en) | 2007-09-04 | 2009-10-27 | Upi Semiconductor Corporation | Voltage generating apparatus and methods |
US7783269B2 (en) | 2007-09-20 | 2010-08-24 | Quantance, Inc. | Power amplifier controller with polar transmitter |
KR20090036670A (en) | 2007-10-10 | 2009-04-15 | 삼성전자주식회사 | Apparatus and method for envelope tracking power amplifier in wireless communication system |
JP5189343B2 (en) | 2007-10-23 | 2013-04-24 | ローム株式会社 | Selector circuit and electronic device using the same |
JP4905344B2 (en) | 2007-12-20 | 2012-03-28 | 富士通株式会社 | Power amplifier |
US7923974B2 (en) | 2008-01-04 | 2011-04-12 | Chil Semiconductor Corporation | Modification of switch activation order in a power supply |
US7782036B1 (en) | 2008-01-07 | 2010-08-24 | National Semiconductor Corporation | Adaptive on-time control for switching regulators |
TWI349410B (en) | 2008-01-08 | 2011-09-21 | Novatek Microelectronics Corp | Change pump circuit |
US7949316B2 (en) | 2008-01-29 | 2011-05-24 | Panasonic Corporation | High-efficiency envelope tracking systems and methods for radio frequency power amplifiers |
US8718582B2 (en) | 2008-02-08 | 2014-05-06 | Qualcomm Incorporated | Multi-mode power amplifiers |
JP5119961B2 (en) | 2008-02-08 | 2013-01-16 | 住友電気工業株式会社 | Envelope tracking power supply circuit and high-frequency amplifier including the same |
US7898268B2 (en) * | 2008-02-15 | 2011-03-01 | Infineon Technologies Ag | Circuit and method for capacitor effective series resistance measurement |
WO2009104420A1 (en) | 2008-02-21 | 2009-08-27 | 株式会社アドバンテスト | Digital modulation signal test device, digital modulator, digital demodulator, and semiconductor device using the devices |
KR101434604B1 (en) | 2008-03-03 | 2014-08-26 | 삼성전자주식회사 | Apparatus and method for bias modulator using zero current switching |
US7928705B2 (en) | 2008-03-12 | 2011-04-19 | Sony Ericsson Mobile Communications Ab | Switched mode voltage converter with low-current mode and methods of performing voltage conversion with low-current mode |
GB2459894A (en) * | 2008-05-09 | 2009-11-11 | Nujira Ltd | Switched supply stage with feedback |
US7915961B1 (en) | 2008-05-13 | 2011-03-29 | National Semiconductor Corporation | Power amplifier multiple stage control for polar modulation circuit |
US7759912B2 (en) | 2008-05-13 | 2010-07-20 | Micrel, Inc. | Adaptive compensation scheme for LC circuits in feedback loops |
US7808323B2 (en) | 2008-05-23 | 2010-10-05 | Panasonic Corporation | High-efficiency envelope tracking systems and methods for radio frequency power amplifiers |
US8369973B2 (en) | 2008-06-19 | 2013-02-05 | Texas Instruments Incorporated | Efficient asynchronous sample rate conversion |
US8823342B2 (en) | 2008-07-07 | 2014-09-02 | Advanced Analogic Technologies Incorporated | Multiple-output dual-polarity DC/DC converters and voltage regulators |
JP4613986B2 (en) | 2008-07-28 | 2011-01-19 | 日本テキサス・インスツルメンツ株式会社 | Switching power supply |
US7990119B2 (en) | 2008-07-29 | 2011-08-02 | Telefonaktiebolaget L M Ericsson (Publ) | Multimode voltage regulator circuit |
US20100027301A1 (en) | 2008-07-31 | 2010-02-04 | Motorola, Inc. | Band-pass current mode control scheme for switching power converters with higher-order output filters |
US8000117B2 (en) | 2008-08-13 | 2011-08-16 | Intersil Americas Inc. | Buck boost function based on a capacitor bootstrap input buck converter |
FI20085808A0 (en) | 2008-08-29 | 2008-08-29 | Nokia Corp | Correcting distortions at power amplifier output |
US20110018626A1 (en) | 2008-10-24 | 2011-01-27 | Advantest Corporation | Quadrature amplitude demodulator and demodulation method |
GB2465552B (en) | 2008-11-18 | 2015-12-09 | Nujira Ltd | Power supply arrangement for multi-stage amplifier |
EP2189870A1 (en) | 2008-11-25 | 2010-05-26 | St Microelectronics S.A. | A switch-mode voltage regulator |
TW201043049A (en) | 2008-12-15 | 2010-12-01 | Mediatek Inc | DC-coupled audio out unit |
US8030995B2 (en) | 2008-12-25 | 2011-10-04 | Hitachi Kokusai Electric Inc. | Power circuit used for an amplifier |
CN102265505B (en) | 2008-12-25 | 2014-04-23 | 日本电气株式会社 | Power amplication device |
EP2214304B1 (en) | 2009-01-30 | 2011-10-12 | Alcatel Lucent | Switch mode assisted linear amplifier for baseband signal amplification |
CN102308474B (en) * | 2009-02-05 | 2015-08-26 | 日本电气株式会社 | Power amplifier and power-magnifying method |
US8138734B2 (en) * | 2009-04-06 | 2012-03-20 | Monolithic Power Systems, Inc. | Accurate current limit for peak current mode DC-DC converter |
US8026765B2 (en) | 2009-04-12 | 2011-09-27 | Roberto Michele Giovannotto | Audio frequency amplifier |
US8093951B1 (en) | 2009-04-14 | 2012-01-10 | Cirrus Logic, Inc. | Pulse-width modulated (PWM) audio power amplifier having output signal magnitude controlled pulse voltage and switching frequency |
US8749213B2 (en) * | 2009-06-09 | 2014-06-10 | Silergy Technology | Mixed mode control for switching regulator with fast transient responses |
JP5365369B2 (en) | 2009-06-26 | 2013-12-11 | 富士通株式会社 | Transmission apparatus, distortion compensation apparatus, and distortion compensation method |
US8081199B2 (en) | 2009-06-26 | 2011-12-20 | Panasonic Corporation | Light emitting element drive apparatus, planar illumination apparatus, and liquid crystal display apparatus |
GB0912745D0 (en) * | 2009-07-22 | 2009-08-26 | Wolfson Microelectronics Plc | Improvements relating to DC-DC converters |
KR20110026065A (en) | 2009-09-07 | 2011-03-15 | 삼성전자주식회사 | Apparatus and method for envelope tracking power amplifier in wireless communication |
JP5343786B2 (en) | 2009-09-18 | 2013-11-13 | ヤマハ株式会社 | Amplification equipment |
US8461815B1 (en) | 2009-10-05 | 2013-06-11 | Huy X Ngo | Fast transient buck regulator with dynamic charge/discharge capability |
TWI418139B (en) | 2009-10-09 | 2013-12-01 | Richtek Technology Corp | Highly efficient class-g amplifier and control method thereof |
US8823343B2 (en) | 2009-12-22 | 2014-09-02 | Yamaha Corporation | Power amplifying circuit, DC-DC converter, peak holding circuit, and output voltage control circuit including the peak holding circuit |
US8548398B2 (en) | 2010-02-01 | 2013-10-01 | Rf Micro Devices, Inc. | Envelope power supply calibration of a multi-mode radio frequency power amplifier |
US8183929B2 (en) | 2010-04-09 | 2012-05-22 | Viasat, Inc. | Multi-chip doherty amplifier with integrated power detection |
US8981848B2 (en) | 2010-04-19 | 2015-03-17 | Rf Micro Devices, Inc. | Programmable delay circuitry |
US8493141B2 (en) | 2010-04-19 | 2013-07-23 | Rf Micro Devices, Inc. | Pseudo-envelope following power management system |
US9099961B2 (en) | 2010-04-19 | 2015-08-04 | Rf Micro Devices, Inc. | Output impedance compensation of a pseudo-envelope follower power management system |
US8542061B2 (en) | 2010-04-20 | 2013-09-24 | Rf Micro Devices, Inc. | Charge pump based power amplifier envelope power supply and bias power supply |
US8706063B2 (en) | 2010-04-20 | 2014-04-22 | Rf Micro Devices, Inc. | PA envelope power supply undershoot compensation |
US8174313B2 (en) | 2010-05-17 | 2012-05-08 | Avago Technologies Wireless Ip (Singapore) Pte. Ltd. | Apparatus and method for controlling power amplifier |
CN101867284B (en) | 2010-05-31 | 2012-11-21 | 华为技术有限公司 | Control method of fast tracking power supply, fast tracking power supply and system |
US8183917B2 (en) | 2010-06-04 | 2012-05-22 | Quantance, Inc. | RF power amplifier circuit with mismatch tolerance |
JP2011259083A (en) | 2010-06-07 | 2011-12-22 | Renesas Electronics Corp | Rf power amplifier and operation method thereof |
US20110298432A1 (en) | 2010-06-07 | 2011-12-08 | Skyworks Solutions, Inc | Apparatus and method for variable voltage function |
US8164391B2 (en) * | 2010-07-28 | 2012-04-24 | Active-Semi, Inc. | Synchronization of multiple high frequency switching power converters in an integrated circuit |
WO2012027039A1 (en) | 2010-08-25 | 2012-03-01 | Rf Micro Devices, Inc. | Multi-mode/multi-band power management system |
US8204456B2 (en) | 2010-09-15 | 2012-06-19 | Fujitsu Semiconductor Limited | Systems and methods for spurious emission cancellation |
EP2432118B1 (en) * | 2010-09-15 | 2012-12-26 | Agence Spatiale Européenne | Radio-frequency power amplifier with fast envelope tracking |
WO2012066659A1 (en) | 2010-11-17 | 2012-05-24 | 株式会社日立製作所 | High-frequency amplifier, and high-frequency module and wireless machine using same |
JP5742186B2 (en) | 2010-11-22 | 2015-07-01 | 富士通株式会社 | Amplifier |
US8674620B2 (en) | 2010-11-30 | 2014-03-18 | Infineon Technologies Ag | Multi channel LED driver |
TWI419448B (en) | 2010-12-02 | 2013-12-11 | Richtek Technology Corp | Power supply circuit with adaptive input selection and method for power supply |
JP5614273B2 (en) | 2010-12-21 | 2014-10-29 | 富士通株式会社 | Amplifier |
US8773102B2 (en) | 2011-01-03 | 2014-07-08 | Eta Semiconductor Inc. | Hysteretic CL power converter |
US8588713B2 (en) | 2011-01-10 | 2013-11-19 | Rf Micro Devices, Inc. | Power management system for multi-carriers transmitter |
US8803605B2 (en) * | 2011-02-01 | 2014-08-12 | Mediatek Singapore Pte. Ltd. | Integrated circuit, wireless communication unit and method for providing a power supply |
US8611402B2 (en) | 2011-02-02 | 2013-12-17 | Rf Micro Devices, Inc. | Fast envelope system calibration |
CN103444076B (en) | 2011-02-07 | 2016-05-04 | 射频小型装置公司 | For the group delay calibration steps of power amplifier envelope-tracking |
US8624760B2 (en) | 2011-02-07 | 2014-01-07 | Rf Micro Devices, Inc. | Apparatuses and methods for rate conversion and fractional delay calculation using a coefficient look up table |
JP5996559B2 (en) | 2011-02-07 | 2016-09-21 | スカイワークス ソリューションズ,インコーポレイテッドSkyworks Solutions,Inc. | Apparatus and method for envelope tracking calibration |
US8576523B2 (en) | 2011-03-14 | 2013-11-05 | Qualcomm Incorporated | Charge pump electrostatic discharge protection |
US8725218B2 (en) | 2011-03-25 | 2014-05-13 | R2 Semiconductor, Inc. | Multimode operation DC-DC converter |
EP4220950A3 (en) | 2011-05-05 | 2023-12-06 | Qorvo US, Inc. | Power management architecture for modulated and constant supply operation |
US8362837B2 (en) | 2011-05-23 | 2013-01-29 | Vyycore Ltd. | System and a method for amplifying a signal by multiple non-linear power amplifiers |
US8638165B2 (en) | 2011-06-06 | 2014-01-28 | Qualcomm Incorporated | Switched-capacitor DC blocking amplifier |
IL213624A (en) | 2011-06-16 | 2016-02-29 | David Leonardo Fleischer | Method and system for boosting the power supply of a power amplifier |
US8626091B2 (en) | 2011-07-15 | 2014-01-07 | Rf Micro Devices, Inc. | Envelope tracking with variable compression |
US8618868B2 (en) | 2011-08-17 | 2013-12-31 | Rf Micro Devices, Inc. | Single charge-pump buck-boost for providing independent voltages |
KR101793733B1 (en) | 2011-10-14 | 2017-11-06 | 삼성전자주식회사 | Apparatus and method for calibration of supply modualtion in transmitter |
US9298198B2 (en) * | 2011-12-28 | 2016-03-29 | Rf Micro Devices, Inc. | Noise reduction for envelope tracking |
US8952753B2 (en) | 2012-02-17 | 2015-02-10 | Quantance, Inc. | Dynamic power supply employing a linear driver and a switching regulator |
US8981839B2 (en) | 2012-06-11 | 2015-03-17 | Rf Micro Devices, Inc. | Power source multiplexer |
US8648657B1 (en) | 2012-08-13 | 2014-02-11 | Broadcom Corporation | Mobile device including a power amplifier with selectable voltage supply |
-
2013
- 2013-07-26 WO PCT/US2013/052277 patent/WO2014018861A1/en active Application Filing
- 2013-07-26 US US13/951,976 patent/US9020451B2/en active Active
- 2013-07-26 CN CN201380039592.1A patent/CN104662792B/en active Active
Patent Citations (6)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20060178119A1 (en) * | 2005-02-09 | 2006-08-10 | Nokia Corporation | Variable bandwidth envelope modulator for use with envelope elimination and restoration transmitter architecture and method |
WO2007149346A2 (en) * | 2006-06-16 | 2007-12-27 | Pulsewave Rf, Inc. | Radio frequency power amplifier and method using a controlled supply |
US20100266066A1 (en) * | 2007-11-05 | 2010-10-21 | Nec Corporation | Power amplifier and radio wave transmitter having the same |
US20120025907A1 (en) * | 2010-07-28 | 2012-02-02 | Korea Advanced Institute Of Science And Technology | Power amplifier |
GB2484475A (en) * | 2010-10-11 | 2012-04-18 | Toshiba Res Europ Ltd | A power supply modulator for an RF amplifier, using a current-output class G amplifier |
US20120154054A1 (en) * | 2010-12-17 | 2012-06-21 | Skyworks Solutions, Inc. | Apparatus and methods for oscillation suppression |
Non-Patent Citations (2)
Title |
---|
PATRICK Y. WU ET AL: "A Two-Phase Switching Hybrid Supply Modulator for RF Power Amplifiers With 9% Efficiency Improvement", IEEE JOURNAL OF SOLID-STATE CIRCUITS, vol. 45, no. 12, 1 December 2010 (2010-12-01), pages 2543 - 2556, XP055081507, ISSN: 0018-9200, DOI: 10.1109/JSSC.2010.2076510 * |
YOUSEFZADEH V ET AL: "Band Separation and Efficiency Optimization in Linear-Assisted Switching Power Amplifiers", POWER ELECTRONICS SPECIALISTS CONFERENCE, 2006. PESC '06. 37TH IEEE JEJU, KOREA 18-22 JUNE 2006, PISCATAWAY, NJ, USA,IEEE, 18 June 2006 (2006-06-18), pages 1 - 7, XP010945542, ISBN: 978-0-7803-9716-3, DOI: 10.1109/PESC.2006.1712188 * |
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US9020451B2 (en) | 2015-04-28 |
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US20140028368A1 (en) | 2014-01-30 |
CN104662792B (en) | 2017-08-08 |
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