CA1277704C - Cross coupled current regulator - Google Patents
Cross coupled current regulatorInfo
- Publication number
- CA1277704C CA1277704C CA000498428A CA498428A CA1277704C CA 1277704 C CA1277704 C CA 1277704C CA 000498428 A CA000498428 A CA 000498428A CA 498428 A CA498428 A CA 498428A CA 1277704 C CA1277704 C CA 1277704C
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- CA
- Canada
- Prior art keywords
- signal
- phase
- integrator
- input
- proportional
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired - Lifetime
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Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
Abstract
Abstract of the Disclosure A current regulator for a variable frequency power supply includes a proportional plus integral circuit that produces a composite control output signal for each phase. To improve the performance of the current regulator at higher frequencies, the composite control output signal for each phase also includes a cross coupled component which is produced by multiplying a d.c.
signal proportional to frequency times an integrator output signal from another phase. The composite control signals are applied to a voltage source inverter which produces the output currents to a load.
signal proportional to frequency times an integrator output signal from another phase. The composite control signals are applied to a voltage source inverter which produces the output currents to a load.
Description
q~'7'7~)~
CROSS COUPLED CURRENT REGULATOR
The field of the invention is variable speed drives for a.c.
motors, and particularly, to alternating current sources which provide polyphase, variable frequency, and variable amplitude currents to motor stator windings.
Prior alternating current sources employ solid state switches to produce pulse waveforms that approximate sinusoidal waveforms suitable for application to the motor's stator windings. These are generally divided into two classes: those which employ current source inverters; and those which employ voltage source inverters. A current source inverter receives a controlled d.c.
"link" current and switches it between the motor stator windings in such a manner as to approximate the application of polyphase sinusoidal currents of the proper frequency and amplitude. Such an inverter is disclosed, for example, in U.S. Patent No.
~,~00,655. A voltage source inverter on the other hand, receives a d.c. voltage and chops it into a series of voltage pulses which are applied to the motor stator windings. The widths of the pulses are modulated such that the resulting currents produced in the motor stator windings have a sinusoidal shape of the desired frequency and amplitude. Such an inverter is shown, for example, in U.S. Patent No. 4,469,997.
Both the current source and the voltage source inverters have their respective advantages and disadvantages which are well-known in the art. As a result, both technolo~ies are employed in co~nercially available motor drive products, with the choice being det~rmined primarily by performance and price con-sideration.
The present invention solves a problem which is inherent in prior voltage s~urce inverters that employ current regulators.
It has long been recognized that at high motor speeds such voltage source inverters do not accurately control sinusoidal motor currents. This has been attributed b~ some to a loss in gain in 7~
voltage source inverters at high speeds, while others relate the problem to the back e.m.f. of the motor. Numerous schemes for solving this problem have been proposed, some of which involve compensation circuitry that is either very complex or that re~uires detailed knowledge about the particular motor being driven.
The inability of such voltage source inverters to accurately produce current waveforms of commanded amplitude and phase is particularly troublesome when such inverters are used in a motor drive that relies on controlling the magnetic flux vector. Such vector control strategies require that the amplitude and phase of the sinusoidal current waveform applied to each stator winding be precisely controlled at all times. Only then will the total magnetic flux have the proper magnitude and direction to produce the desired motor torque and speed.
The present invention relates to variable frequency a.c.
power supplies which employ voltage source inverters, and particu-larly, to a means for improving the current regulating capability of such inverters at high frequencies. The present invention includes means for producing an error signal by subtracting an a.c. current feedback signal from an a.c. current command signal and means for producing a composite control signal for a voltage source inverter by adding a signal component proportional to the error signal to a signal component proportional to the integral of the error signal and to a cross-coupled component which increases in amplitude as a unction of frequency. The cross-coupled component is produced by a multiplier which connects to receive a signal component from another phase of the current regulator and to receive a signal proportional to the frequency of the a.c. current command signal.
The invention enables one to accurately regulate the a.c.
current produced by a vol-tage source inverter. The current regulator of the present invention eliminates errors which are
CROSS COUPLED CURRENT REGULATOR
The field of the invention is variable speed drives for a.c.
motors, and particularly, to alternating current sources which provide polyphase, variable frequency, and variable amplitude currents to motor stator windings.
Prior alternating current sources employ solid state switches to produce pulse waveforms that approximate sinusoidal waveforms suitable for application to the motor's stator windings. These are generally divided into two classes: those which employ current source inverters; and those which employ voltage source inverters. A current source inverter receives a controlled d.c.
"link" current and switches it between the motor stator windings in such a manner as to approximate the application of polyphase sinusoidal currents of the proper frequency and amplitude. Such an inverter is disclosed, for example, in U.S. Patent No.
~,~00,655. A voltage source inverter on the other hand, receives a d.c. voltage and chops it into a series of voltage pulses which are applied to the motor stator windings. The widths of the pulses are modulated such that the resulting currents produced in the motor stator windings have a sinusoidal shape of the desired frequency and amplitude. Such an inverter is shown, for example, in U.S. Patent No. 4,469,997.
Both the current source and the voltage source inverters have their respective advantages and disadvantages which are well-known in the art. As a result, both technolo~ies are employed in co~nercially available motor drive products, with the choice being det~rmined primarily by performance and price con-sideration.
The present invention solves a problem which is inherent in prior voltage s~urce inverters that employ current regulators.
It has long been recognized that at high motor speeds such voltage source inverters do not accurately control sinusoidal motor currents. This has been attributed b~ some to a loss in gain in 7~
voltage source inverters at high speeds, while others relate the problem to the back e.m.f. of the motor. Numerous schemes for solving this problem have been proposed, some of which involve compensation circuitry that is either very complex or that re~uires detailed knowledge about the particular motor being driven.
The inability of such voltage source inverters to accurately produce current waveforms of commanded amplitude and phase is particularly troublesome when such inverters are used in a motor drive that relies on controlling the magnetic flux vector. Such vector control strategies require that the amplitude and phase of the sinusoidal current waveform applied to each stator winding be precisely controlled at all times. Only then will the total magnetic flux have the proper magnitude and direction to produce the desired motor torque and speed.
The present invention relates to variable frequency a.c.
power supplies which employ voltage source inverters, and particu-larly, to a means for improving the current regulating capability of such inverters at high frequencies. The present invention includes means for producing an error signal by subtracting an a.c. current feedback signal from an a.c. current command signal and means for producing a composite control signal for a voltage source inverter by adding a signal component proportional to the error signal to a signal component proportional to the integral of the error signal and to a cross-coupled component which increases in amplitude as a unction of frequency. The cross-coupled component is produced by a multiplier which connects to receive a signal component from another phase of the current regulator and to receive a signal proportional to the frequency of the a.c. current command signal.
The invention enables one to accurately regulate the a.c.
current produced by a vol-tage source inverter. The current regulator of the present invention eliminates errors which are
-2-7 ~ 7 0 L~
introduced into the current regulator by the conventional propor-tional plus integral (PI) compensation circuit that is uniformly employed. It has been discovered that such conventional PI com-pensation circuits are not accurate when regulating a.c. currents.
This inaccuracy increases as a function of the a.c. frequency of the currents being regulated, and this accounts for the difficul-ties which are encountered when current regulated voltage source inverters are employed to drive electric motors. Unlike prior circuits, the present invention provides accurate current regula-tion under steady-state conditions at any operating frequency.
The invention also enables one to provide current regulation for a voltage source inverter which is economical to build and operate. The improved current regulator includes only a few passive components and integrated circuits in addition to those normally employed in a conventional PI current regulator.
The invention enables one to improve the operation of poly-phase motor drive systems which employ vector control strategies to control motor speed, torque or magnetic flu~. The present invention may be employed to control the magnitude of n separate a.c. phase currents in response to two orthogonal a.c. current co}nmand signals. The magnitude and phase of the resulting motor currents accurately reflect the commanded currents at all operat-ing frequencies.
In drawings which illustrate the embodiments of the invention, Fig. 1 i9 a block diagram of a current regulated voltage source invert~r driving a two-phase load;
Fig. 2 is a block diagram of a current regulated voltage source inverter driving a three-phase load;
Fig. 3 is a functional diagram of a current regulator which employs the present invention;
Fig. ~ is an electrical schematic diagram of the current regulation of Fig. 3; and ~1.'2~7~7~;'0~
Fig. 5 are graphic representations of currents and voltages which appear at various points in the system of Fig. 1.
Referring particularly to Fig. 1, a load 1 is supplied with two-phase a~c. power by a voltage source inverter 2. The load 1 may be any one of a number of devices such as a synchronous motor, induction motor, electrostatic precipitator, induction heating unlt or corona treater. The voltage source inverter 2 is a well-known circuit which produces pulse width modulated voltage pulses on the lines 3 and ~ in response to control signals on lines 5 and 6. As shown in Fig. 5, these voltage pulses have a constant amplitude (V), but their widths are modulatad such that the currents, i~ and id flowing in the respective lines 3 and 4 are substantially sinusoidal in shape. ~he amplitude, frequency and phase of these output currents iq and id are determined ~y the amplitude, frequency and phase of the control signals on lines 5 and ~. Voltage source inverters such as those described in U.S. Patent Nos. 4,469,997; 3,830,003 and 3,700,987 may be employed ~or this purpose.
The output currents iq and id are precisely controlled by a current regulator 7 which produces the control signals on the inverter input lines 5 and 6. The current regulator 7 receives two sinusoidal current command signals, iq = Iq sin ~t and id = Id sin (~t-90), which are compared with sinusoidal current feedback signals i~ and id that are received from current sensors through respective lines 8 and 9. It is the function of the current regulator 7 to produce sinusoidal control signals on the lines 5 and 6 which will drive the voltage source inverter 2 in such a manner as to cause the respective ~eedback signals`i~
and id to equal t~e current command signa's iq and id. There are numerous well known current regulators which purport to perorm this function, but in all but the most complex circuiks, these prior current regulators fail to perform accuratel~ over a wide . -4-7 70L~
range of frequencies. As will be discussed in more detail below, as part of the solution to this problem the current regulator 7 of the present invention employs a d.c. input signal (~) on line 10 which has a magnitude proportional to the frequency of the command currents iq and id.
Referrin~ particularly to Fig. 2, a current regulated voltage source inverter system may also be employed to drive a three-phase load 20. A ~oltage source inverter 21 similar to that used in the two-phase system may be employed, however, it is driven by three sinusoidal control signals on lines 22-24 and it produces three output currents ia, ib and ic. The three-phase output currents (ia, ib~ ic) have the same magnitude and frequency, but they are displaced 120 degrees in phase.
The same regulator 7 may be employed in this three-phase system, but phase conversions must be made. More specifically, the two-phase current regulator control signals which are output on lines 5 and 6 must be converted to the equivalent three-phase si~nals on lines 22-24. This conversion is per~ormed by a 2-phase to-3-phase converter ci~cuit 25 of well-known construc-tion. For example, a circuit such as that described in "Controland Simulation of a Current Fed Linear Inductor Machine" by B. K.
Bose and Thomas Lipo published in IEEE-IAS Confere_ce Record, pp. 876-883, 1978, may be employed for this purpose. Conversely, the three output currents ia, ib and ic which are fed back through lines 26-28 are converted to two-phase feedback signals iq and id by a 3-phase-to-2-phase converter circuit 29. A circ~it such as that described in "Control Methods for Good Dynamic Performance Induction Motor Drives Based on Current and Voltage as Measured Quantities", by Robert Joetten and Gerhard Maeder and published 30 in IEEE-IAS Transactions, IA-l9, No. 3, Ma~y/June 1983, may be employed for this purpose.
It should be apparent to those skilled in the art that tha current regulator of the present invention may be employed in a 70~
wide variety of applications to control a.c. loads having any number of phases. Also, the current command signals i~ and i as well as the speed signal ~, may be produced by any one of a number of well-known control circuits. The specific construc-tion of the control circuit will depend on the nature of the loadand on the particular control st~ategy which is being implemented.
A number of different control circuits for a.c. motors are dis-closed in U.S. Patent Nos. 4,506,321 and 4,266,176.
The current regulator of the present in~ention provides a combination of proportional control action, integral control action, and cross coupled speed compensation action. The ~Iql~
phase and the "d" phase are handled in the same manner, and the corresponding functional blocks and circuit elements in each phase of the current regulator have been given the same reference number.
Referring particularly to Fig. 3, the sinusoidal current feedback signal iq is subtracted from the sinusoidal current command signal iq = ~q sin ~t at a summing point 50q. The resulting error signal teq) produces a proportional control ~ignal through a proportional block 51q and an integral control signal through blocks 52q and 53q. These two control signals are added together at summing point 54q to produce the composite control signal on line 5.
This composite control signal also includes a crosscoupled component which increases in magnitude at higher frequencies (~). It is produced by a multiplier block 55~ that recei~es a cross-coupled signal from the integrator block 53d o the other phase. The amplitude of this crosscoupled signal is modulated by a second input to the multiplier 55q which is proportional to frequency. This second input is the d.c. frequency signal (~) on the line 10. The cross-coupled component which is output by ~ '~t77-7~)~
the multiplier 55q is added at a summing point 56q which connects to the input of the integrator 53~.
Under steady state conditions the error signals e~ and ed should be zero. The composite control signals on the lines 5 and 6, however, must be waveforms which cause the voltage source inverter 21 to produce output currents corresponding to the commanded currents iq and id. Under most operating conditions these composite control signals are substantially sinusoidal in shape as shown in Fig. 5. These steady state sinusoidal composita control signals are produced by the cross-coupling of the present invention. More specifically, under steady state conditions the cross connections between the two phases ~orm an oscillator which operates at the commanded frequency. The amplitude of the cross-coupled compon~nt produced by this oscillator is proportional to frequency. If for any reason an error signal develops in either phase, the input to the affected integrator 53q or 53d will at that instant include both the cross-coupled component and an error somponent (KIe). The resulting composite control signals on the corresponding output line 5 or 6 will include the integral of these two components and a component proportional to the error signal (Kpe). This composite control signal will force the error signals eq and e~ to zero through a coordinated control between the phases to insure balanced control even at higher frequencies.
Referring particularly to Figs. 3 and 4, the preferred embodiment of the invention is constructed using passive compon-ents and standard, commarcially available integrated circuits.
The summing point SOq, for example, is implemented with an opera-tional amplifier lOOq which receives the current command signal i~ and current ~eedback signal i~ at its inverting input. The values of resistors R2 are tha same to provide unity gain at the summing point 50~. Similarly, the summing point 54q at the output is implemented with an operational amplifier lOlq. The ~ -7-1 ~'777(~
values of resistors Rp and ~ provide the gain (Kp) for the proportional block 51q, and the values of resistors R3 and R4 are the same to provide unitv gain for the integral component.
The integral block 53g is implemented with an operational amplifier 102g having a feedback capacitor C connected between its output and its inverting input. An input resistor RI, con-nects to the same inverting input and its value relative to the value of capacitor C provides the integral gain (kI). An inverter having unity gain is formed by an operational ampli~ier 103q and associated resistors Rl. This inverter insures that the signal produced by the integral block 53q has the same sign as the signal produced by the proportional block 51q.
The inverting input to the operational amplifier 102q also forms the summing point 56~ which receives the cross coupled signal ~rom the multiplier 55q. The multiplier 55q employs a commercially available integrated circuit manu~actured by Motorola, Inc. and sold as part number MC 1595L. Its output connects to the sunlming point 56q through a resistor having a value RI/C. This provides a unity gain for the cross coupled component.
The multiplier 55q is described in more detail starting at page 6-83 in the book "Linear Integrated Circuits", published by Motorola, Inc. in 1979. One of its inputs connects to the line 10 to receive the frequency signal ~, and its other input Z5 connects to the output of the integrator in the other phase. The multiplier 55d is connected in a similar manner, but it connects to receive the inversion o the integrator output from the opera-tional amplifier 103q.
An oscillator is formed by these cross connections. More specifically, a loop is formed by the multiplier 55q, integrator 102q, inverter 103q, multiplier 55d and integrator lO~d. The phase shifts around this loop total 360 degrees which causes it 7~7()~
to oscillate at the frequency w. The amplitude of the cross coupled signals produced by this oscillator is determined by the requency w. The rnagnitude of the frequency signal w is set to produce maximum possible signals at the outputs of the multipliers 55q and 55d when the highest operating frequency is reached.
While the preferred embodiment of the invention employs two phases (q and d) which are in quadrature with each other, current regulators having other numbers of phases may also be constructed.
It is only necessary that the cross coupled signals have the proper phase and amplitude relationship. Referring to Fig. 3, for example, the g-phase composite control signal on the line 5 lags the d-phase composite control signal on line 6 by ninety degrees. The cross coupled signal from integrator block 53q is shifted 180 degrees in phase by the inversion of the frequency signal ~ into the multiplier 55d~ and it is delayed another 90 degrees by the irltegrator block 53d to bring it into phase with the d phase composite control signal on line ~. Conversely, the cross coupled signal from the output o integrator block 53d is merely delayed 90 degrees by the integrator block 53q to bring it into phase with the q-phase composite control signal on line 6. It should be apparent to those skilled in the art that with three or more phases cross coupled signals would be received from each of the other phases, and that the vector sum of these cross coupled signals should be in phase with the subject phase and have unity gain as described above.
introduced into the current regulator by the conventional propor-tional plus integral (PI) compensation circuit that is uniformly employed. It has been discovered that such conventional PI com-pensation circuits are not accurate when regulating a.c. currents.
This inaccuracy increases as a function of the a.c. frequency of the currents being regulated, and this accounts for the difficul-ties which are encountered when current regulated voltage source inverters are employed to drive electric motors. Unlike prior circuits, the present invention provides accurate current regula-tion under steady-state conditions at any operating frequency.
The invention also enables one to provide current regulation for a voltage source inverter which is economical to build and operate. The improved current regulator includes only a few passive components and integrated circuits in addition to those normally employed in a conventional PI current regulator.
The invention enables one to improve the operation of poly-phase motor drive systems which employ vector control strategies to control motor speed, torque or magnetic flu~. The present invention may be employed to control the magnitude of n separate a.c. phase currents in response to two orthogonal a.c. current co}nmand signals. The magnitude and phase of the resulting motor currents accurately reflect the commanded currents at all operat-ing frequencies.
In drawings which illustrate the embodiments of the invention, Fig. 1 i9 a block diagram of a current regulated voltage source invert~r driving a two-phase load;
Fig. 2 is a block diagram of a current regulated voltage source inverter driving a three-phase load;
Fig. 3 is a functional diagram of a current regulator which employs the present invention;
Fig. ~ is an electrical schematic diagram of the current regulation of Fig. 3; and ~1.'2~7~7~;'0~
Fig. 5 are graphic representations of currents and voltages which appear at various points in the system of Fig. 1.
Referring particularly to Fig. 1, a load 1 is supplied with two-phase a~c. power by a voltage source inverter 2. The load 1 may be any one of a number of devices such as a synchronous motor, induction motor, electrostatic precipitator, induction heating unlt or corona treater. The voltage source inverter 2 is a well-known circuit which produces pulse width modulated voltage pulses on the lines 3 and ~ in response to control signals on lines 5 and 6. As shown in Fig. 5, these voltage pulses have a constant amplitude (V), but their widths are modulatad such that the currents, i~ and id flowing in the respective lines 3 and 4 are substantially sinusoidal in shape. ~he amplitude, frequency and phase of these output currents iq and id are determined ~y the amplitude, frequency and phase of the control signals on lines 5 and ~. Voltage source inverters such as those described in U.S. Patent Nos. 4,469,997; 3,830,003 and 3,700,987 may be employed ~or this purpose.
The output currents iq and id are precisely controlled by a current regulator 7 which produces the control signals on the inverter input lines 5 and 6. The current regulator 7 receives two sinusoidal current command signals, iq = Iq sin ~t and id = Id sin (~t-90), which are compared with sinusoidal current feedback signals i~ and id that are received from current sensors through respective lines 8 and 9. It is the function of the current regulator 7 to produce sinusoidal control signals on the lines 5 and 6 which will drive the voltage source inverter 2 in such a manner as to cause the respective ~eedback signals`i~
and id to equal t~e current command signa's iq and id. There are numerous well known current regulators which purport to perorm this function, but in all but the most complex circuiks, these prior current regulators fail to perform accuratel~ over a wide . -4-7 70L~
range of frequencies. As will be discussed in more detail below, as part of the solution to this problem the current regulator 7 of the present invention employs a d.c. input signal (~) on line 10 which has a magnitude proportional to the frequency of the command currents iq and id.
Referrin~ particularly to Fig. 2, a current regulated voltage source inverter system may also be employed to drive a three-phase load 20. A ~oltage source inverter 21 similar to that used in the two-phase system may be employed, however, it is driven by three sinusoidal control signals on lines 22-24 and it produces three output currents ia, ib and ic. The three-phase output currents (ia, ib~ ic) have the same magnitude and frequency, but they are displaced 120 degrees in phase.
The same regulator 7 may be employed in this three-phase system, but phase conversions must be made. More specifically, the two-phase current regulator control signals which are output on lines 5 and 6 must be converted to the equivalent three-phase si~nals on lines 22-24. This conversion is per~ormed by a 2-phase to-3-phase converter ci~cuit 25 of well-known construc-tion. For example, a circuit such as that described in "Controland Simulation of a Current Fed Linear Inductor Machine" by B. K.
Bose and Thomas Lipo published in IEEE-IAS Confere_ce Record, pp. 876-883, 1978, may be employed for this purpose. Conversely, the three output currents ia, ib and ic which are fed back through lines 26-28 are converted to two-phase feedback signals iq and id by a 3-phase-to-2-phase converter circuit 29. A circ~it such as that described in "Control Methods for Good Dynamic Performance Induction Motor Drives Based on Current and Voltage as Measured Quantities", by Robert Joetten and Gerhard Maeder and published 30 in IEEE-IAS Transactions, IA-l9, No. 3, Ma~y/June 1983, may be employed for this purpose.
It should be apparent to those skilled in the art that tha current regulator of the present invention may be employed in a 70~
wide variety of applications to control a.c. loads having any number of phases. Also, the current command signals i~ and i as well as the speed signal ~, may be produced by any one of a number of well-known control circuits. The specific construc-tion of the control circuit will depend on the nature of the loadand on the particular control st~ategy which is being implemented.
A number of different control circuits for a.c. motors are dis-closed in U.S. Patent Nos. 4,506,321 and 4,266,176.
The current regulator of the present in~ention provides a combination of proportional control action, integral control action, and cross coupled speed compensation action. The ~Iql~
phase and the "d" phase are handled in the same manner, and the corresponding functional blocks and circuit elements in each phase of the current regulator have been given the same reference number.
Referring particularly to Fig. 3, the sinusoidal current feedback signal iq is subtracted from the sinusoidal current command signal iq = ~q sin ~t at a summing point 50q. The resulting error signal teq) produces a proportional control ~ignal through a proportional block 51q and an integral control signal through blocks 52q and 53q. These two control signals are added together at summing point 54q to produce the composite control signal on line 5.
This composite control signal also includes a crosscoupled component which increases in magnitude at higher frequencies (~). It is produced by a multiplier block 55~ that recei~es a cross-coupled signal from the integrator block 53d o the other phase. The amplitude of this crosscoupled signal is modulated by a second input to the multiplier 55q which is proportional to frequency. This second input is the d.c. frequency signal (~) on the line 10. The cross-coupled component which is output by ~ '~t77-7~)~
the multiplier 55q is added at a summing point 56q which connects to the input of the integrator 53~.
Under steady state conditions the error signals e~ and ed should be zero. The composite control signals on the lines 5 and 6, however, must be waveforms which cause the voltage source inverter 21 to produce output currents corresponding to the commanded currents iq and id. Under most operating conditions these composite control signals are substantially sinusoidal in shape as shown in Fig. 5. These steady state sinusoidal composita control signals are produced by the cross-coupling of the present invention. More specifically, under steady state conditions the cross connections between the two phases ~orm an oscillator which operates at the commanded frequency. The amplitude of the cross-coupled compon~nt produced by this oscillator is proportional to frequency. If for any reason an error signal develops in either phase, the input to the affected integrator 53q or 53d will at that instant include both the cross-coupled component and an error somponent (KIe). The resulting composite control signals on the corresponding output line 5 or 6 will include the integral of these two components and a component proportional to the error signal (Kpe). This composite control signal will force the error signals eq and e~ to zero through a coordinated control between the phases to insure balanced control even at higher frequencies.
Referring particularly to Figs. 3 and 4, the preferred embodiment of the invention is constructed using passive compon-ents and standard, commarcially available integrated circuits.
The summing point SOq, for example, is implemented with an opera-tional amplifier lOOq which receives the current command signal i~ and current ~eedback signal i~ at its inverting input. The values of resistors R2 are tha same to provide unity gain at the summing point 50~. Similarly, the summing point 54q at the output is implemented with an operational amplifier lOlq. The ~ -7-1 ~'777(~
values of resistors Rp and ~ provide the gain (Kp) for the proportional block 51q, and the values of resistors R3 and R4 are the same to provide unitv gain for the integral component.
The integral block 53g is implemented with an operational amplifier 102g having a feedback capacitor C connected between its output and its inverting input. An input resistor RI, con-nects to the same inverting input and its value relative to the value of capacitor C provides the integral gain (kI). An inverter having unity gain is formed by an operational ampli~ier 103q and associated resistors Rl. This inverter insures that the signal produced by the integral block 53q has the same sign as the signal produced by the proportional block 51q.
The inverting input to the operational amplifier 102q also forms the summing point 56~ which receives the cross coupled signal ~rom the multiplier 55q. The multiplier 55q employs a commercially available integrated circuit manu~actured by Motorola, Inc. and sold as part number MC 1595L. Its output connects to the sunlming point 56q through a resistor having a value RI/C. This provides a unity gain for the cross coupled component.
The multiplier 55q is described in more detail starting at page 6-83 in the book "Linear Integrated Circuits", published by Motorola, Inc. in 1979. One of its inputs connects to the line 10 to receive the frequency signal ~, and its other input Z5 connects to the output of the integrator in the other phase. The multiplier 55d is connected in a similar manner, but it connects to receive the inversion o the integrator output from the opera-tional amplifier 103q.
An oscillator is formed by these cross connections. More specifically, a loop is formed by the multiplier 55q, integrator 102q, inverter 103q, multiplier 55d and integrator lO~d. The phase shifts around this loop total 360 degrees which causes it 7~7()~
to oscillate at the frequency w. The amplitude of the cross coupled signals produced by this oscillator is determined by the requency w. The rnagnitude of the frequency signal w is set to produce maximum possible signals at the outputs of the multipliers 55q and 55d when the highest operating frequency is reached.
While the preferred embodiment of the invention employs two phases (q and d) which are in quadrature with each other, current regulators having other numbers of phases may also be constructed.
It is only necessary that the cross coupled signals have the proper phase and amplitude relationship. Referring to Fig. 3, for example, the g-phase composite control signal on the line 5 lags the d-phase composite control signal on line 6 by ninety degrees. The cross coupled signal from integrator block 53q is shifted 180 degrees in phase by the inversion of the frequency signal ~ into the multiplier 55d~ and it is delayed another 90 degrees by the irltegrator block 53d to bring it into phase with the d phase composite control signal on line ~. Conversely, the cross coupled signal from the output o integrator block 53d is merely delayed 90 degrees by the integrator block 53q to bring it into phase with the q-phase composite control signal on line 6. It should be apparent to those skilled in the art that with three or more phases cross coupled signals would be received from each of the other phases, and that the vector sum of these cross coupled signals should be in phase with the subject phase and have unity gain as described above.
Claims (5)
1. In a variable frequency power supply which receives a plurality of sinusoidal current command signals and produces sinusoidal output currents to a load, a polyphase current regula-tor for receiving the plurality of sinusoidal current command signals and producing composite control signals, each current regulator phase comprising:
first summing means for receiving at one input one of said sinusoidal current command signals and receiving at a second input a sinusoidal current feedback signal indicative of sinus-oidal output current supplied to said load, the first summing means being operable to produce an error signal which is indica-tive of the difference between the sinusoidal current command signal and the sinusoidal current feedback signal;
second summing means for receiving at one input a signal proportional to said error signal and for receiving at a second input a cross coupled signal, the second summing means being operable to produce a summed signal which is proportional to the sum of the signals applied to its two inputs;
an integrator having an input connected to receive the summed signal from the second summing means and being operable to produce an integrator output signal which is the integral of the summed signal applied to its input;
third summing means for receiving at one input the integrator output signal and for receiving at a second input a signal propor-tional to said error signal, the third summing means being oper-able to produce one of the composite control signal which is proportional to the sum of the signals applied to its inputs; and means for producing said cross coupled signal which includes:
a) a multiplier having its output connected to the second summing means, b) means for coupling to one input on the multiplier a signal which is proportional in magnitude to the frequency of said one sinusoidal current command, and c) means for coupling to another input on the multiplier a signal received from another of said current regulator phases.
first summing means for receiving at one input one of said sinusoidal current command signals and receiving at a second input a sinusoidal current feedback signal indicative of sinus-oidal output current supplied to said load, the first summing means being operable to produce an error signal which is indica-tive of the difference between the sinusoidal current command signal and the sinusoidal current feedback signal;
second summing means for receiving at one input a signal proportional to said error signal and for receiving at a second input a cross coupled signal, the second summing means being operable to produce a summed signal which is proportional to the sum of the signals applied to its two inputs;
an integrator having an input connected to receive the summed signal from the second summing means and being operable to produce an integrator output signal which is the integral of the summed signal applied to its input;
third summing means for receiving at one input the integrator output signal and for receiving at a second input a signal propor-tional to said error signal, the third summing means being oper-able to produce one of the composite control signal which is proportional to the sum of the signals applied to its inputs; and means for producing said cross coupled signal which includes:
a) a multiplier having its output connected to the second summing means, b) means for coupling to one input on the multiplier a signal which is proportional in magnitude to the frequency of said one sinusoidal current command, and c) means for coupling to another input on the multiplier a signal received from another of said current regulator phases.
2. The current regulator as recited in claim 1 in which the signal received from said another current regulator phase is proportional to the integrator output signal produced by an integrator in said another current regulator phase.
3. The current regulator as recited in claim 2 in which the current regulator has two phases and the two sinusoidal current command signals are phase displaced ninety degrees with respect to each other.
4. In a current regulator having a pair of phases, each phase providing a composite control signal to a voltage source inverter in response to a sinusoidal current command signal, each phase comprising:
first summing means for producing an error signal equal to the difference between the sinusoidal current command signal and a current feedback signal;
integrator means coupled to the first summing means for producing an integrator output signal which is proportional to the integral of the error signal;
second summing means coupled to the first summing means and the integrator means for producing a composite control signal which is proportional to the error signal plus the integrator output signal;
means for producing a frequency signal proportional to the frequency of the sinusoidal current command signal; and cross-coupling means for producing a cross-coupled signal which is applied to the integrator means and which is proportional to an integrator output signal from the other phase which is modulated in amplitude by the magnitude of the frequency signal.
first summing means for producing an error signal equal to the difference between the sinusoidal current command signal and a current feedback signal;
integrator means coupled to the first summing means for producing an integrator output signal which is proportional to the integral of the error signal;
second summing means coupled to the first summing means and the integrator means for producing a composite control signal which is proportional to the error signal plus the integrator output signal;
means for producing a frequency signal proportional to the frequency of the sinusoidal current command signal; and cross-coupling means for producing a cross-coupled signal which is applied to the integrator means and which is proportional to an integrator output signal from the other phase which is modulated in amplitude by the magnitude of the frequency signal.
5. In a current regulator having a pair of phases, each phase including an integrator which receives at its input an error signal formed by summing a sinusoidal current command and a current feedback signal, and which produces a signal at its output that is employed to control the current in a load, the improvement therein comprising:
an oscillator loop for applying cross-coupled signals to each phase, and being formed by coupling the output of the inte-grator in each phase to the input of the integrator in the other phase, said oscillator loop including means connected to receive a signal indicative of the frequency of the sinusoidal current command and for controlling the magnitude of the cross-coupled signals as a function of said frequency.
an oscillator loop for applying cross-coupled signals to each phase, and being formed by coupling the output of the inte-grator in each phase to the input of the integrator in the other phase, said oscillator loop including means connected to receive a signal indicative of the frequency of the sinusoidal current command and for controlling the magnitude of the cross-coupled signals as a function of said frequency.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US73565985A | 1985-05-20 | 1985-05-20 | |
| US735,659 | 1985-05-20 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1277704C true CA1277704C (en) | 1990-12-11 |
Family
ID=24956681
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000498428A Expired - Lifetime CA1277704C (en) | 1985-05-20 | 1985-12-23 | Cross coupled current regulator |
Country Status (5)
| Country | Link |
|---|---|
| US (2) | US4706012A (en) |
| EP (1) | EP0202603B1 (en) |
| JP (1) | JPH0834693B2 (en) |
| CA (1) | CA1277704C (en) |
| DE (1) | DE3688342T2 (en) |
Families Citing this family (23)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| EP0274716A1 (en) * | 1987-01-09 | 1988-07-20 | Siemens Aktiengesellschaft | Method and device to determine the flux vector of an induction machine |
| US5140248A (en) * | 1987-12-23 | 1992-08-18 | Allen-Bradley Company, Inc. | Open loop motor control with both voltage and current regulation |
| US4958117A (en) * | 1989-09-29 | 1990-09-18 | Allen-Bradley Company, Inc. | Frequency control based on sensing voltage fed to an induction motor |
| US5122730A (en) * | 1990-01-02 | 1992-06-16 | The United States Of America As Represented By The Secretary Of The Air Force | Voltage divider for a wide band domino effect high voltage regulator |
| US5032771A (en) * | 1990-08-09 | 1991-07-16 | Allen-Bradley Company, Inc. | Slip control based on sensing voltage fed to an induction motor |
| US5136228A (en) * | 1990-12-11 | 1992-08-04 | Kabushiki Kaisha Meidensha | Induction motor vector control |
| US5274323A (en) * | 1991-10-31 | 1993-12-28 | Linear Technology Corporation | Control circuit for low dropout regulator |
| US5220262A (en) * | 1992-02-25 | 1993-06-15 | Cincinnati Milacron, Inc. | Method and apparatus for reducing cross-coupled movement through the structural dynamics of a computer numerically controlled machine |
| US5619114A (en) * | 1995-03-20 | 1997-04-08 | Allen-Bradley Company, Inc. | Signal averager for use with motor controller |
| US5721474A (en) * | 1995-08-11 | 1998-02-24 | Samsung Electronics Co., Ltd. | Method and apparatus for preventing excessive current flow in a motor |
| US6018225A (en) * | 1998-02-09 | 2000-01-25 | Allen-Bradley Company, Llc | Method and apparatus for reconnecting a rotating motor to a motor drive |
| US6459230B1 (en) | 2001-02-13 | 2002-10-01 | Rockwell Automation Technologies, Inc. | Method and system for measuring a parameter of motor operation |
| WO2003073185A2 (en) | 2002-02-28 | 2003-09-04 | Zetacon Corporation | Predictive control system and method |
| US7809053B2 (en) * | 2002-04-04 | 2010-10-05 | Infineon Technologies Ag | Device and method for control scaling and quantization of soft output values from an equalizer |
| DE10214908A1 (en) * | 2002-04-04 | 2003-11-13 | Infineon Technologies Ag | Device and method for the controlled scaling and quantization of soft output values of an equalizer |
| US6965212B1 (en) | 2004-11-30 | 2005-11-15 | Honeywell International Inc. | Method and apparatus for field weakening control in an AC motor drive system |
| ITVA20070011A1 (en) * | 2007-01-17 | 2008-07-18 | St Microelectronics Srl | METHOD AND ESTIMATION DEVICE OF THE VALUES OF TWO CIRCULATING CURRENTS IN THE SAME INSTANT IN THEIR PHASES OF A POLYPHASE LOAD PILOTED IN SWM MODE |
| US7894169B2 (en) * | 2008-06-02 | 2011-02-22 | Rockwell Automation Technologies, Inc. | High resistance ground protection employing AC drive characteristics |
| US8154228B2 (en) * | 2009-06-10 | 2012-04-10 | Kollmorgen Corporation | Dynamic braking for electric motors |
| JP5488880B2 (en) * | 2009-09-25 | 2014-05-14 | 富士電機株式会社 | Winding induction machine controller |
| EP2451062B1 (en) * | 2010-11-08 | 2018-10-24 | Nxp B.V. | PFC with multi-channel error feedback |
| TWI556555B (en) * | 2013-02-07 | 2016-11-01 | Variable flow control module with suppression of harmonic action | |
| EP2881811B1 (en) * | 2013-12-06 | 2018-08-01 | General Electric Technology GmbH | A control apparatus for a voltage source converter comprising a redundant control system of two controllers |
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| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| DE2833542C2 (en) * | 1978-07-31 | 1980-09-25 | Siemens Ag, 1000 Berlin Und 8000 Muenchen | Rotary field machine drive, consisting of a converter-fed rotary field machine, in particular a synchronous machine and a converter control for the self-clocked, in particular field-oriented operation of this machine, with two identical AC voltage integrators and a method for operating the rotary field machine drive |
| JPS56162995A (en) * | 1980-05-15 | 1981-12-15 | Fuji Electric Co Ltd | Load compensating circuit for variable frequency inverter |
| JPS5815492A (en) * | 1981-07-20 | 1983-01-28 | Hitachi Ltd | Control of pulse width control transducer |
| US4382275A (en) * | 1981-12-07 | 1983-05-03 | Sundstrand Corporation | PWM Inverter circuit |
| JPS59103589A (en) * | 1982-12-01 | 1984-06-15 | Hitachi Ltd | Controller for power converter |
| US4509003A (en) * | 1983-03-10 | 1985-04-02 | Kabushiki Kaisha Meidensha | Vector control method and system for an induction motor |
| JPS59169369A (en) * | 1983-03-16 | 1984-09-25 | Toshiba Corp | Ac current controller |
-
1985
- 1985-12-23 CA CA000498428A patent/CA1277704C/en not_active Expired - Lifetime
-
1986
- 1986-01-14 JP JP61006173A patent/JPH0834693B2/en not_active Expired - Lifetime
- 1986-05-14 US US06/863,015 patent/US4706012A/en not_active Expired - Lifetime
- 1986-05-15 EP EP86106586A patent/EP0202603B1/en not_active Expired - Lifetime
- 1986-05-15 DE DE86106586T patent/DE3688342T2/en not_active Expired - Lifetime
- 1986-10-27 US US06/922,775 patent/US4680695A/en not_active Expired - Lifetime
Also Published As
| Publication number | Publication date |
|---|---|
| US4680695A (en) | 1987-07-14 |
| EP0202603A3 (en) | 1987-10-14 |
| JPH0834693B2 (en) | 1996-03-29 |
| JPS61266076A (en) | 1986-11-25 |
| EP0202603B1 (en) | 1993-04-28 |
| US4706012A (en) | 1987-11-10 |
| DE3688342D1 (en) | 1993-06-03 |
| EP0202603A2 (en) | 1986-11-26 |
| DE3688342T2 (en) | 1993-10-28 |
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