CA1276242C - Wide band inductive transdermal power and data link - Google Patents
Wide band inductive transdermal power and data linkInfo
- Publication number
- CA1276242C CA1276242C CA000505519A CA505519A CA1276242C CA 1276242 C CA1276242 C CA 1276242C CA 000505519 A CA000505519 A CA 000505519A CA 505519 A CA505519 A CA 505519A CA 1276242 C CA1276242 C CA 1276242C
- Authority
- CA
- Canada
- Prior art keywords
- coil
- link
- coils
- coupling
- gain
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired - Fee Related
Links
Classifications
-
- A—HUMAN NECESSITIES
- A61—MEDICAL OR VETERINARY SCIENCE; HYGIENE
- A61N—ELECTROTHERAPY; MAGNETOTHERAPY; RADIATION THERAPY; ULTRASOUND THERAPY
- A61N1/00—Electrotherapy; Circuits therefor
- A61N1/18—Applying electric currents by contact electrodes
- A61N1/32—Applying electric currents by contact electrodes alternating or intermittent currents
- A61N1/36—Applying electric currents by contact electrodes alternating or intermittent currents for stimulation
- A61N1/372—Arrangements in connection with the implantation of stimulators
- A61N1/37211—Means for communicating with stimulators
- A61N1/37217—Means for communicating with stimulators characterised by the communication link, e.g. acoustic or tactile
- A61N1/37223—Circuits for electromagnetic coupling
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y10—TECHNICAL SUBJECTS COVERED BY FORMER USPC
- Y10S—TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y10S128/00—Surgery
- Y10S128/903—Radio telemetry
Abstract
ABSTRACT
The present invention utilizes a pair of coils, one beneath the skin and one outside the skin, each connected to a capacitor. The values of the capacitor and coil inductance are selected to provide a stag-ger tuned link. That is, the values of the components are selected to place the pole of one filter above the operating frequency, and the pole of the other filter below the operating frequency of the link. The poles will move as the coupling coefficient changes, desensitizing the link to the coupling so that the coils can be mis-aligned in any manner with little effect on the output.
A fully active driver or class D amplifier is preferably utilized to effectively drive the link with a square wave signal. This signal can be modulated to convey data information to the implanted receiv-er coil beneath the skin.
The present invention utilizes a pair of coils, one beneath the skin and one outside the skin, each connected to a capacitor. The values of the capacitor and coil inductance are selected to provide a stag-ger tuned link. That is, the values of the components are selected to place the pole of one filter above the operating frequency, and the pole of the other filter below the operating frequency of the link. The poles will move as the coupling coefficient changes, desensitizing the link to the coupling so that the coils can be mis-aligned in any manner with little effect on the output.
A fully active driver or class D amplifier is preferably utilized to effectively drive the link with a square wave signal. This signal can be modulated to convey data information to the implanted receiv-er coil beneath the skin.
Description
A WIDE: BAND IND~7CTIVE TRANSDERMAL
POI~ER AND DATA LINK
The present invention is directed generally to the field of medical electronics, and more particu]arly to a method and apparatus for 5 conveying power and data through the skin of a subject without breaking the skin.
With the increase in use of implanted electronic devices beneath human skin, it has been increasingly desirable to be able to convey power and/or data to the implanted device without puncturing the 10 skin. A typical approach previously followed was to use a percu-taneous plug which could deliver power, but must break the skin to reach the stimulator. Breaching the skin provides a potential path for infection to enter, risking the safety of the individual as well as the implant. Another alternative is to implant a battery. t~nfor-15 tunately, many applications require more amp hours than a batterycan provide without frequent reimplantation.
The use of an inductive link has neither of these limitations since it leaves the skin intact and can draw from an outside power source.
For medical applications, an inductive link is usually composed of 20 two pancake shaped coils, one implanted in the body and one out-side the body. These coils form a transformer through which ener-gy can be sent. Inductive links are already used to deliver power and information into implanted stimulators. Examples of this can be found in the Ph.D. Dissertation of M. Soma, "Design and Fabrica-25 tion of an Implantable Multi-channel Neural Stimulator" Ph.D. Dis-sertation June, 1980; the Ph.D. Dissertation of T. Gheewala, "A
CMOS Implantable Auditory Prosthesis tlsing Integrated Circuit Technology," Stanford t~niversity, March 1976, and the article by $
` ~276242 E.S. Hochmair, IEEE ~ransactions on Biomedical Engineering, Vol, BME-31, No.2, pp.l77-186, February 1984, entitled, "System Optimi-zation for Improved Accuracy in Transcutaneous Signal and Power Transmission, n 5 However, inductive links as considered in these prior efforts have their own set of difficulties. In the past, these ]inks have had prob]ems providing efficiency, bandwidth, and insensitivity to misa]ignment .
It is an objective of the present invention to provide a new and 10 improved inductive link.
It is another objective of this invention to provide such a link whose gain is relatively insensitive to coupling variations between the coils which occur with relative movement between the coils.
Yet another objective of the present invention is to provide an 15 inductive link which has a relatively wide bandwidth.
Efficiency is one of the major problems to be attacked in designing such coup]ing systems. Efficiency is a prob]em because the freedom of the imp]anted person is limited by the portability of the trans-mitter power source. If the transdermal link is inefficient, an 20 implant of even modest power demand may require a large amount of power to the transmitter. If the system requires an excessive amount of power, the individual must carry a large battery pack.
For example, the systems described in the above-referenced Ph.D.
theses required a battery pack about the size of a brick which 25 needed recharging every eight hours.
Therefore, it is an objective of the present invention to provide a system which is small and efficient enough for a person to wear freely and use in everyday activities.
Inductive links have been inefficient because designs to compensate 30 for coil misalignment have been inefficient. ~he coils can move relative to one another. As they move, their gain (voltage gain, current gain, transconductance gain, or transimpedance gain, depending on the input and output circuitry) changes, since their coupling depends on their relative positions. Many applications require a stable internal power supply, and therefore, must regulate the output of the receiver coil.
If the gain variation is too large, the most of the received power is not used in the stimulator, but wasted in the voltage regulator coupled to the receiving coil \
The most commonly used approach to solving this problem, called the geometric approach, sacrifices the coupling magnitude in exchange for reduction in the coupling variation. It uses an external coil which iæ larger than the implanted coil. As long as the small coil remains within the perimeter of the larger coil, both coils will share roughly the same number of flux lines, and the coupling will remain fairly constant. The coils can move laterally and even tilt somewhat with only a minor effect on their gain. ~his approach is adopted in one or more of the above references.
Unfortunately, in addition to being inefficient, the geometric approach is still sensitive to changes in coil separation; for a small separation (near field), the coupling drops proportionally with separation. If the coil separation doubles, then the gain is cut in half. This approach, therefore, only partially deæensitizes the link gain to the coil's position.
Secondly, in exchange for partially controlling the gain, the geometric approach sacrifices the magnitude of the coupling, which raises the lo~ses in the primary coil. Since , .. .
~276~:42 3a these coils share fewer flux llnes than two equally sized coils, their typical coupling coefficient is lower than their equally sized counterparts. In fact, their coupling is much lower since their shared area diminishes as the square of the lateral misalignment tolerance, i.e., shared area -n(larger radius-alignment tolerance) . Lower coupling requires a higher current in the primary coil to provide the same output from the ~2762A2 receiver coil; and higher current wastes more power due to the I2R
losses of the transmitter coil.
The earlier links also had another problem, i . e ., poor bandwidth .
These links were usually single tuned (transmitter coil only) or 5 double tuned (both coils) to the carrier frequency 60 that they cou]d be driven by a semi-passive RF transmitter (Class C, for example) for higher efficiency. Semi-passive transmitters need a high Q tuned circuit to operate efficiently. ~1nfortunately, the higher the Q, the smaller the band width. The previous Stanford 10 stimulators used a separate ultrasonic data link to obtain a wide band data path, but that link proved much less tolerant to misalign-ment than the inductive link, and unreliable in use due to acoustic impedance mismatches. Clearly, an efficient wide band power and data link would solve some difficult problems.
15 In summary, the new approach described in the present invention utilizes a pair of coils, one beneath the skin and one outside the skin, each connected to a capacitor. The values of the capacitor and coil inductance are selected to provide a stagger-tuned link. That is, the values of the components are selected to place the pole of 20 one filter above the operating frequency, and the pole of the other filter below the operating frequency of the link. The poles will move as the coupling coefficient changes, desensitizing the link to the coupling so that the coils can be misaligned in any manner with little effect on the output.
25 The present invention includes a number of approaches which differ from obvious uses of stagger tuning or any prior art approaches.
The problem is that when the coils move, the coupling coefficient changes and so does the ratio of input to output. The present invention applies stagger tuning to desensitize the link going into 30 the coupling coefficient rather than broaden the band width as is normally done in stagger-tuned filters.
Second, the gain is stabilized due to an approach which takes into account and relies on the interaction of two components; typically, ~1.27624~
gain is stabilized by isolating components from affecting one another.
In a further departure from the approaches of the prior art, a fully active driver or class D amplifier is preferably utilized to effectively drive the link with a square wave signal.
This signal can be modulated to convey data information to the implanted receiver coil beneath the skin.
In accordance with a broad aspect of the invention there is provided an inductive link for conveying power and/or data through the skin of a user comprising a first external coil, a second medically implantable coil, the coils being movable relative to one another, the mutual inductive coupling of said coils changing with their position relative to one another, and a means for stagger tuning said first and second coils to maintain a gain which is relatively independent of the coil's mutual coupling at a selected frequency or frequency band of interest.
Other advantageq of the present invention will become apparent from the following detailed disclosure given with reference to the accompanying drawings wherein:
Figure 1 illustrates the various coil misalignments dealt with by the present invention, Figures 2A through 2D show the four possible comblnations of stagger-tuned links, Figures 3A to 3C illustrate the effective result of a coupling compensated link voltage device using the circuit combination shown in Figure 2A, Figure 4 is a second graph illustrating the effective relationship of gain versus coupling and load impedance at a given B
~L276242 Sa operating frequency of 20 MHz, and Figure 5 is a detailed schematic of a Class D
transmitter useful in driving the link of the present invention.
The "Voltage Driven:Voltage Out" link of Figure 2A will illustrate the approach of the present invention to gain control.
The voltage gain across an untuned or singly tuned two-coil system varies linearly with the coupling coefficient (assuming the load resistance is large compared to the coil reactance), so if the coupling increases by a factor of three, it will cause the voltage gain to triple. Thls lax galn control is generally unacceptable because it usually wastes most of the received power in the voltage regulatlon.
Flgure 2A is a schematic of the transmitter and receiver elements ~276242 of an inductive link (Lt and Lr), the tuning elements (or=l/~Fr and ~t=l/-/L~ the transmitter loss and receiver load elements (Rt and Rr) and the transmitter source and receiver voltages. This circuit reduces the coupling dependent gain 5 variation by tuning the coils to two different frequencies; one above and one below the operating frequency. If these coils' resonances are placed optimally, their effective poles will shift as the mutual coupling of the coils changes. The gain wi]] be compensated as a resu]t .
10 The nvo]tage driven:voltage out" link will help illustrate the new approach to gain control. Figure 3 is a plot of the voltage gain versus frequency and coupling for a link that was designed to ope-rate at 20 MHz. The numbers on the curves represent the values of coupling coefficient; 2 means K=O . 2, 3 means K=O . 3, and so on .
15 The link is stagger-tuned; i.e., the component values are selected as shown so that one pole is placed above the operating frequency (20 MHz) and one pole below. The poles move as the coupling coef-ficient changes. If they are placed properly, their positions move to compensate for the changing coupling. To illustrate, in Figure 3 20 as the coupling increases, the poles move away from the operating frequency, Increased coupling would normally raise the gain at 20 MHz, but the higher coupling has moved the po]es away from the 20 MHz operating frequency, therefore lowering the gain at the operat-ing frequency. The net result, for this example, is that even 25 though the coupling coefficient changes by 250%, the gain changes by les~: that ~10%.
Since this approach is based on coupling, not geometry, it not on]y corrects for latera] displacement and tilt, as did the prior art geometric approach, but it also handles coil separation, which the 30 geometric solution did not, Also notice that the gain changes smoothly around the operating frequency (20 MHz), giving a stable gain over a relatively large frequency band. Classically, ~tagger tuning has been used to increase the bandwldth of a system without excessively loading the tuned circuits. As the example of Figure 3 shows, the new approach to gain control has carried over the attractive feature of a wide bandwidth. For many applications, the additional bandwidth can simplify the design. For example, power and data links are combined into one inductive link by sending power on the carrier and the data by frequency-modulating tFM) the carrier as discussed in U.S. patent No. 4,592,359 of Galbraith, et al.
entitled "An Eight Channel Implantable Neural Stimulator".
This example has clearly demonstrated a wide link gain which i8 coupling-insensitive. With these capabilities, the power lost across the internal voltage regulators of an implanted stimulator can be reduced by taking advantage of the stable gain.
By uslng equal sized coils, lnstead of the mismatched coil sizes of the geometrlc approach, the average coupling can be improved which lowers the tranæmitter coil I2R losses. The coupling-compensated llnk not only allows improved efflciency, but it also provldes a large bandwldth, something that the previous approaches have sacriflced for efflclency.
To take advantage of the efflclency afforded by the - stagger-tuned llnk's stable galn, the link needs a transmitter that can efflclently handle a variable load impedance whlle malntalnlng a constant drlving voltage or current. Seml-passive drlvers, such as class C or claæs E, are not practical because they lose efficiency if the load impedance is not resonant at the operating frequency, a fundamental lncompatibility with stagger tuning. A class D transmitter, on the other hand, can drive these loads efficiently, and can supply a constant driving source so J
~L276242 7a ~ 61051-1952 that the link output voltage, or current, remains stable.
A class D transmitter, on the other hand, can drive these loads efficlently, because it can supply a constant source which is independent of the load. It simply switches the input of the link between the two terminals of the power supply. Reactive loads and ~2762A2 load variations, due to changing coup]ing, should not affect its output level.
This driver also has a theoretical maximum efficiency of 100%, mean-ing that its efficiency is not bounded by a sub-optimum theoretical 5 limit, an advantage in any design. The driver's loss is proportional to the ratio of the voltage dropped across its switches to the supply voltage. It is fairly simple to make this ratio quite small, and in practice, the design tends to become more efficient as the output power level is raised. In fact, the efficiency improves when it is 10 needed most, when the transmitter is delivering the maximum power.
The class D driver, therefore, meets the requirements for driving the stagger tuned link; it can efficiently drive a variable load with a stable source.
A class D driver has another advantage due to its square wave out-15 put. The Fourier Series of a square wave has a fundamental sine wave whose amplitude is 4/7r times the amplitude of the square wave.
The stagger tuned link passes the fundamental and attenuates the harmonics, meaning that the class D driver provides the added advantage of another 27% increase in gain. Other design considera-20 tions may trade off this additional gain to further advantage. Forexample, Figure 4 shows the theoretical and experimental gain versus coupling for the 20 MHz link described above. It has a correction factor of 0. 745 which means that the theoretical gain is multiplied by 0 . 745 to match the experimental results . The differ-25 ence between theory and practice comes from operating the coilsnear their self-resonant frequencies. The parasitic capacitance of the transmitter coil acts with the tuning capacitance to form a voltage divider between the transmitter and the inductor. It lowers the amount of voltage reaching the tuned circuit. In this example, 30 the additional square wave gain (27~o) was used to replenish the gain lost to the self resonance. A typical class D driver usable with the coils described above is shown in Figure 5.
~2762A2 Now that the concept behind the gain control and its drive require-ments have been described and examples shown, it is appropriate to discuss a design procedure, including some of the de~ign tradeoffs.
The following procedure was used to create a stagger tuned power 5 and data link; a "voltage driveD:voltage out" linlc. The goals for this procedure were:
l) to minimize the variation in output voltage, due to chang-ing load and coupling, over a band of frequencies;
POI~ER AND DATA LINK
The present invention is directed generally to the field of medical electronics, and more particu]arly to a method and apparatus for 5 conveying power and data through the skin of a subject without breaking the skin.
With the increase in use of implanted electronic devices beneath human skin, it has been increasingly desirable to be able to convey power and/or data to the implanted device without puncturing the 10 skin. A typical approach previously followed was to use a percu-taneous plug which could deliver power, but must break the skin to reach the stimulator. Breaching the skin provides a potential path for infection to enter, risking the safety of the individual as well as the implant. Another alternative is to implant a battery. t~nfor-15 tunately, many applications require more amp hours than a batterycan provide without frequent reimplantation.
The use of an inductive link has neither of these limitations since it leaves the skin intact and can draw from an outside power source.
For medical applications, an inductive link is usually composed of 20 two pancake shaped coils, one implanted in the body and one out-side the body. These coils form a transformer through which ener-gy can be sent. Inductive links are already used to deliver power and information into implanted stimulators. Examples of this can be found in the Ph.D. Dissertation of M. Soma, "Design and Fabrica-25 tion of an Implantable Multi-channel Neural Stimulator" Ph.D. Dis-sertation June, 1980; the Ph.D. Dissertation of T. Gheewala, "A
CMOS Implantable Auditory Prosthesis tlsing Integrated Circuit Technology," Stanford t~niversity, March 1976, and the article by $
` ~276242 E.S. Hochmair, IEEE ~ransactions on Biomedical Engineering, Vol, BME-31, No.2, pp.l77-186, February 1984, entitled, "System Optimi-zation for Improved Accuracy in Transcutaneous Signal and Power Transmission, n 5 However, inductive links as considered in these prior efforts have their own set of difficulties. In the past, these ]inks have had prob]ems providing efficiency, bandwidth, and insensitivity to misa]ignment .
It is an objective of the present invention to provide a new and 10 improved inductive link.
It is another objective of this invention to provide such a link whose gain is relatively insensitive to coupling variations between the coils which occur with relative movement between the coils.
Yet another objective of the present invention is to provide an 15 inductive link which has a relatively wide bandwidth.
Efficiency is one of the major problems to be attacked in designing such coup]ing systems. Efficiency is a prob]em because the freedom of the imp]anted person is limited by the portability of the trans-mitter power source. If the transdermal link is inefficient, an 20 implant of even modest power demand may require a large amount of power to the transmitter. If the system requires an excessive amount of power, the individual must carry a large battery pack.
For example, the systems described in the above-referenced Ph.D.
theses required a battery pack about the size of a brick which 25 needed recharging every eight hours.
Therefore, it is an objective of the present invention to provide a system which is small and efficient enough for a person to wear freely and use in everyday activities.
Inductive links have been inefficient because designs to compensate 30 for coil misalignment have been inefficient. ~he coils can move relative to one another. As they move, their gain (voltage gain, current gain, transconductance gain, or transimpedance gain, depending on the input and output circuitry) changes, since their coupling depends on their relative positions. Many applications require a stable internal power supply, and therefore, must regulate the output of the receiver coil.
If the gain variation is too large, the most of the received power is not used in the stimulator, but wasted in the voltage regulator coupled to the receiving coil \
The most commonly used approach to solving this problem, called the geometric approach, sacrifices the coupling magnitude in exchange for reduction in the coupling variation. It uses an external coil which iæ larger than the implanted coil. As long as the small coil remains within the perimeter of the larger coil, both coils will share roughly the same number of flux lines, and the coupling will remain fairly constant. The coils can move laterally and even tilt somewhat with only a minor effect on their gain. ~his approach is adopted in one or more of the above references.
Unfortunately, in addition to being inefficient, the geometric approach is still sensitive to changes in coil separation; for a small separation (near field), the coupling drops proportionally with separation. If the coil separation doubles, then the gain is cut in half. This approach, therefore, only partially deæensitizes the link gain to the coil's position.
Secondly, in exchange for partially controlling the gain, the geometric approach sacrifices the magnitude of the coupling, which raises the lo~ses in the primary coil. Since , .. .
~276~:42 3a these coils share fewer flux llnes than two equally sized coils, their typical coupling coefficient is lower than their equally sized counterparts. In fact, their coupling is much lower since their shared area diminishes as the square of the lateral misalignment tolerance, i.e., shared area -n(larger radius-alignment tolerance) . Lower coupling requires a higher current in the primary coil to provide the same output from the ~2762A2 receiver coil; and higher current wastes more power due to the I2R
losses of the transmitter coil.
The earlier links also had another problem, i . e ., poor bandwidth .
These links were usually single tuned (transmitter coil only) or 5 double tuned (both coils) to the carrier frequency 60 that they cou]d be driven by a semi-passive RF transmitter (Class C, for example) for higher efficiency. Semi-passive transmitters need a high Q tuned circuit to operate efficiently. ~1nfortunately, the higher the Q, the smaller the band width. The previous Stanford 10 stimulators used a separate ultrasonic data link to obtain a wide band data path, but that link proved much less tolerant to misalign-ment than the inductive link, and unreliable in use due to acoustic impedance mismatches. Clearly, an efficient wide band power and data link would solve some difficult problems.
15 In summary, the new approach described in the present invention utilizes a pair of coils, one beneath the skin and one outside the skin, each connected to a capacitor. The values of the capacitor and coil inductance are selected to provide a stagger-tuned link. That is, the values of the components are selected to place the pole of 20 one filter above the operating frequency, and the pole of the other filter below the operating frequency of the link. The poles will move as the coupling coefficient changes, desensitizing the link to the coupling so that the coils can be misaligned in any manner with little effect on the output.
25 The present invention includes a number of approaches which differ from obvious uses of stagger tuning or any prior art approaches.
The problem is that when the coils move, the coupling coefficient changes and so does the ratio of input to output. The present invention applies stagger tuning to desensitize the link going into 30 the coupling coefficient rather than broaden the band width as is normally done in stagger-tuned filters.
Second, the gain is stabilized due to an approach which takes into account and relies on the interaction of two components; typically, ~1.27624~
gain is stabilized by isolating components from affecting one another.
In a further departure from the approaches of the prior art, a fully active driver or class D amplifier is preferably utilized to effectively drive the link with a square wave signal.
This signal can be modulated to convey data information to the implanted receiver coil beneath the skin.
In accordance with a broad aspect of the invention there is provided an inductive link for conveying power and/or data through the skin of a user comprising a first external coil, a second medically implantable coil, the coils being movable relative to one another, the mutual inductive coupling of said coils changing with their position relative to one another, and a means for stagger tuning said first and second coils to maintain a gain which is relatively independent of the coil's mutual coupling at a selected frequency or frequency band of interest.
Other advantageq of the present invention will become apparent from the following detailed disclosure given with reference to the accompanying drawings wherein:
Figure 1 illustrates the various coil misalignments dealt with by the present invention, Figures 2A through 2D show the four possible comblnations of stagger-tuned links, Figures 3A to 3C illustrate the effective result of a coupling compensated link voltage device using the circuit combination shown in Figure 2A, Figure 4 is a second graph illustrating the effective relationship of gain versus coupling and load impedance at a given B
~L276242 Sa operating frequency of 20 MHz, and Figure 5 is a detailed schematic of a Class D
transmitter useful in driving the link of the present invention.
The "Voltage Driven:Voltage Out" link of Figure 2A will illustrate the approach of the present invention to gain control.
The voltage gain across an untuned or singly tuned two-coil system varies linearly with the coupling coefficient (assuming the load resistance is large compared to the coil reactance), so if the coupling increases by a factor of three, it will cause the voltage gain to triple. Thls lax galn control is generally unacceptable because it usually wastes most of the received power in the voltage regulatlon.
Flgure 2A is a schematic of the transmitter and receiver elements ~276242 of an inductive link (Lt and Lr), the tuning elements (or=l/~Fr and ~t=l/-/L~ the transmitter loss and receiver load elements (Rt and Rr) and the transmitter source and receiver voltages. This circuit reduces the coupling dependent gain 5 variation by tuning the coils to two different frequencies; one above and one below the operating frequency. If these coils' resonances are placed optimally, their effective poles will shift as the mutual coupling of the coils changes. The gain wi]] be compensated as a resu]t .
10 The nvo]tage driven:voltage out" link will help illustrate the new approach to gain control. Figure 3 is a plot of the voltage gain versus frequency and coupling for a link that was designed to ope-rate at 20 MHz. The numbers on the curves represent the values of coupling coefficient; 2 means K=O . 2, 3 means K=O . 3, and so on .
15 The link is stagger-tuned; i.e., the component values are selected as shown so that one pole is placed above the operating frequency (20 MHz) and one pole below. The poles move as the coupling coef-ficient changes. If they are placed properly, their positions move to compensate for the changing coupling. To illustrate, in Figure 3 20 as the coupling increases, the poles move away from the operating frequency, Increased coupling would normally raise the gain at 20 MHz, but the higher coupling has moved the po]es away from the 20 MHz operating frequency, therefore lowering the gain at the operat-ing frequency. The net result, for this example, is that even 25 though the coupling coefficient changes by 250%, the gain changes by les~: that ~10%.
Since this approach is based on coupling, not geometry, it not on]y corrects for latera] displacement and tilt, as did the prior art geometric approach, but it also handles coil separation, which the 30 geometric solution did not, Also notice that the gain changes smoothly around the operating frequency (20 MHz), giving a stable gain over a relatively large frequency band. Classically, ~tagger tuning has been used to increase the bandwldth of a system without excessively loading the tuned circuits. As the example of Figure 3 shows, the new approach to gain control has carried over the attractive feature of a wide bandwidth. For many applications, the additional bandwidth can simplify the design. For example, power and data links are combined into one inductive link by sending power on the carrier and the data by frequency-modulating tFM) the carrier as discussed in U.S. patent No. 4,592,359 of Galbraith, et al.
entitled "An Eight Channel Implantable Neural Stimulator".
This example has clearly demonstrated a wide link gain which i8 coupling-insensitive. With these capabilities, the power lost across the internal voltage regulators of an implanted stimulator can be reduced by taking advantage of the stable gain.
By uslng equal sized coils, lnstead of the mismatched coil sizes of the geometrlc approach, the average coupling can be improved which lowers the tranæmitter coil I2R losses. The coupling-compensated llnk not only allows improved efflciency, but it also provldes a large bandwldth, something that the previous approaches have sacriflced for efflclency.
To take advantage of the efflclency afforded by the - stagger-tuned llnk's stable galn, the link needs a transmitter that can efflclently handle a variable load impedance whlle malntalnlng a constant drlving voltage or current. Seml-passive drlvers, such as class C or claæs E, are not practical because they lose efficiency if the load impedance is not resonant at the operating frequency, a fundamental lncompatibility with stagger tuning. A class D transmitter, on the other hand, can drive these loads efficiently, and can supply a constant driving source so J
~L276242 7a ~ 61051-1952 that the link output voltage, or current, remains stable.
A class D transmitter, on the other hand, can drive these loads efficlently, because it can supply a constant source which is independent of the load. It simply switches the input of the link between the two terminals of the power supply. Reactive loads and ~2762A2 load variations, due to changing coup]ing, should not affect its output level.
This driver also has a theoretical maximum efficiency of 100%, mean-ing that its efficiency is not bounded by a sub-optimum theoretical 5 limit, an advantage in any design. The driver's loss is proportional to the ratio of the voltage dropped across its switches to the supply voltage. It is fairly simple to make this ratio quite small, and in practice, the design tends to become more efficient as the output power level is raised. In fact, the efficiency improves when it is 10 needed most, when the transmitter is delivering the maximum power.
The class D driver, therefore, meets the requirements for driving the stagger tuned link; it can efficiently drive a variable load with a stable source.
A class D driver has another advantage due to its square wave out-15 put. The Fourier Series of a square wave has a fundamental sine wave whose amplitude is 4/7r times the amplitude of the square wave.
The stagger tuned link passes the fundamental and attenuates the harmonics, meaning that the class D driver provides the added advantage of another 27% increase in gain. Other design considera-20 tions may trade off this additional gain to further advantage. Forexample, Figure 4 shows the theoretical and experimental gain versus coupling for the 20 MHz link described above. It has a correction factor of 0. 745 which means that the theoretical gain is multiplied by 0 . 745 to match the experimental results . The differ-25 ence between theory and practice comes from operating the coilsnear their self-resonant frequencies. The parasitic capacitance of the transmitter coil acts with the tuning capacitance to form a voltage divider between the transmitter and the inductor. It lowers the amount of voltage reaching the tuned circuit. In this example, 30 the additional square wave gain (27~o) was used to replenish the gain lost to the self resonance. A typical class D driver usable with the coils described above is shown in Figure 5.
~2762A2 Now that the concept behind the gain control and its drive require-ments have been described and examples shown, it is appropriate to discuss a design procedure, including some of the de~ign tradeoffs.
The following procedure was used to create a stagger tuned power 5 and data link; a "voltage driveD:voltage out" linlc. The goals for this procedure were:
l) to minimize the variation in output voltage, due to chang-ing load and coupling, over a band of frequencies;
2) to keep a reasonably high input impedance so a transmitter 10 would not need to supply excessively large driving currents;
3) to make the input impedance inductive so that a class D
voltage driver would switch cleanly; and 4 ) to exceed a minimum required voltage gain .
This procedure initially determines whether a design is feasible. If 15 so, then it use6 an iterative process to meet the design goals.
The procedure begins by ~pecifying the type of link, coupling range, load range, and operating frequency. These choices may be limited by size constraints, FCC regulations, or power requirements.
The equation below is an approximation which applies to all of the 20 link combinations. It gives the ratio of the magnitude of the mini-mum gain to the maximum gain (G) in terms of a ratio of the mini-mum and maximum coupling coefficients (Kmin and Kmax). Note that O<K . <K < and O<G<I.
- mm- max- - --G Igainmin I ~ 2~Kmax /Kmin 2~ Igain 1 1+K /K
max max min The coupling range (Kmin and Kmax) is usually found experimental-ly . If the gain variation is too large ( G is too small), then the coupling range needs to be reduced (preferably by raising the min-imum value of coupling which has the added advantages of lowering 30 the transmitter coil losses by reducing the transmitter current, and also improving the load insensitivity of the gain by lowering the link output impedance). One possible solution is to use larger coil ~27~42 diameters. A larger diameter wil] reduce the coupling range as ]ong as the alignment tolerance is held constant. Another fiolution is to use a ferrite backing on either or both coils. The ferrite will compress the magnetic field around the coils and raise their mutual 5 coupling.
Since the gain is still somewhat dependent on coupling, it is neces-sary to find the values of coupling which produce the minimum and maximum values of gain. The minimum gain occurs, by design, at the minimum and maximum values of coupling (K i and ~ ), The 10 maximum gain occurs at the optimum value of coupling given by equation Kopt = KminKmax From here on, the procedure tends to be iterative. Next, the min-imum inductance of each of the two coils is chosen. The receiver coil inductance must provide a reasonable Q (for example, greater 15 than lO) for the loaded receiver tank circuit. If the Q is too low, the output voltage will be sensitive to changes in the load. Also, to maintain a reasonable overall efficiency, the transmitter coi]
inductance should provide a large enough input impedance so that the transmitter will not have to supply an excessive current into the 20 link (generally, raising the transmitter coil's inductance raises the link's input impedance).
The next step is to set the minimum gain and reduce the gain vari-ation over a band of frequencies. A ratio of receiver resonance to 25 the operating frequency (~rl~) is chosen. (Remember, if the link is driven by a class D transmitter, (~r/~) should be chosen so that the input impedance looks inductive, which allows the transmitter to provide efficient switching . ) This ratio along with the resistances, coupling range and coil inductances, fixes the ratio of the transmit-30 ter resonance to the operating frequency (~t/~). The equationswhich specify ~tl~ are different for each type of link and are given below .
~762A2 Next, plots are made of the gain versus frequency for the six com-binations of` minimum and maximum load (Rmjn,RmaX); and minimum, optimum, and maximum values of coupling (kmin,Kopt,lCmaX). The required gain is obtained by varying the turns ratio (inductance of 5 the coils and adjusting ~r/~. For example, if the gain is too small, ~' then the turns ratio should be changed or ~r/~ moved cl4~urc to unity. If the gain variation, due to loading, is too large over the frequency band, then the receiver coil Q should be increased. If the spread is lop-sided (more variation on one side of the band than 10 on the other), then ~r/~ needs to be slightly adjusted until the gain is acceptably uniform about the operating frequency.
There are four combinations of link gain; voltage gain, current gain, transconductance gain, and transimpedance gain. The link gain equation for all four combinations has the form F(K)= AK
-(B K~+C)+j(DK2+E) where k is the coupling coefficient (OCK~ anci A, B, C, D, and E
are variables dependent on the link combination.
J(k)=~ k~ ~/~
where X=B 2~D2 Y=2BC I 2DE
Z=C 2~E2 25 Summarizing for the general section: the minimum gain occurs at k i and kmaX, while the maximum gain occurs at kopt ~mi n~max and is given by f (kopt) ~
The ratio of the minimum gain to the maximum gain is ~27~242 Ikmax 2~
Ge min k max , kmin Therefore, the amount of gain variation (G) and the coupling values where the minimum and maximum gains occur (f in=f (kmin)=f (k ), 10 and fmax=f (kopt) ) are known by simply specifying the coupling range .
Figure 2A is a circuit diagram of a stagger-tuned link which is driven by a voltage source and whose output appears as a voltage source. This arrangement is the type used by the preferred em-15 bodiment. For this circuit the general parameters are ~cl), C~~ Q~, D= ~a 25 E=
u~= 41C~
u~,s= 1 4C, C!, = L~4 ~ ~4 ~276242 Note that ~opt = Z/X = (C2+E2) / (B2+D2) . So substituting andsimplifying give 5 ko~ ZI ~ t~-Solving for the ratio of the operating frequency to the transmitter resonance ~2/~l~t and simplifying gives ~ l-H~V Q~
where H=~
Note the one restriction Qt ~rl Therefore, given the minimum and maximum coupling coefficient, the 20 coil Q's, the operating frequency, and the resonant frequency of the receiver, equation gives the transmitter resonance that minimizes the sensitivity of the gain to the coupling variations. The other three possible stagger-tuned links shown in Fig. 3 follow an equiva-lent approach.
25 Further details of an exemplary embodiment will be given below.
link type: "voltage driven: voltage out"
* coupling range: 0.2~K<0.5 * load range: IKS2'R<3Kn operating frequency: 20 MHz ~ frequency range: 19MHz<f<21MHz transmitter type: class D voltage source * acceptable gain variation across the bandwidth, inc]uding the load variation: 0. 5 drop from the maximum * minimum gain: =2. 0 The ratio of minimum to maximum voltage gain i3 approxima~ely 0 . 90, 5 which is well within the goal of 0. 5 . Next, two identical coi]s, with ferrite backings for high coupling, are chosen. They are 1. 961Jh with a series resistance of 5 . 8Q at 20 MHz . These numbers give a transmitter tank Q of about 40 and a receiver tank Q, including the o l,cn load, of about four.
Next ~ r/w is chosen greater than one so that the input appears inductive for the class D transmitter. Let ~r/~1. 1.
Figure 3B is a p]ot of the gain versus frequency for the six combi-nations of coupling and load, using the equations in the appendix 15 for a "voltage driven:voltage out" link.
Figure 3C is an expanded view of the galn around the operating frequency (20 M~z). The maximurn gain deviation occurs at the high end of the frequency range (Z1 MHz). The ratio of minimum to maximum gain is 0 . 7 which exceeds the design goal of 0. 5 . The 20 minimum gain is about 1.9 which i~ only 5% below the goal of Z, well within the tolerancec of most components, and probably acceptable.
Remember that the receiver Q wa rather low, only 4. The depen-dence of the output voltage on the load is the result of a voltage division between the link output impedance and the effective load 25 impedance. If the link impedance is signi~icant compared to the load, then the output voltage will be load dependent. The curves labeled 1 and 4 on Fi~ure 3C illustrate this point. These curves represent the gain for the minimu~n and maximum load3 at the lowest value o coupling. This link has its highest output irr~pedance when 30 the coils are coupling the least. Over the design band (19 ME1~
21MHz), the large t gain spread occurs at 21 MHz and is bounded by curve~ 1 and 4. Here, the output impedance of the linlc (equiv-alent to about 800 ohms) is significant compared to the load, so the ~L276242 output voltage is somewhat load sensitive. Two ways of reducing the output impedance are: one, raise the coupling coefficient (use ferrite backings and larger coil diameters); and two, make sure that the Q of the loaded receiver coil is reasonably high (adjust the 5 receiver coil inductance).
As mentioned before, this link places some requirements on the driver. Figure 5 shows a preferred embodiment of a transmitter design used to drive the link. This circuit is a class D FM voltage driver with several features for portable, efficient operation. Its 10 frequency is fairly insensitive to changes in the power supply volt-age, so that it can operate from transistor radio batteries without regulation. Its frequency deviation is controlled by a CMOS com-patible input which interfaces easily to portable digital logic. It also uses core drivers in the output stage to rapidly and efficiently 15 switch the high output currents.
The stagger-tuned link, in this example, has a minimum input im-pedance of about 47û, a fairly small load for the transmitter to drive. An efficiency of 35% allows the link to operate for a full day (16 hours) on two nine volt transistor radio batteries. This effici-20 ency provideg a major reduction in power consumption and batterysize over previous Stanford stimulators (a worst case power con-sumption of 140mW instead of 400mW, and two nine volt batteries instead of a brick-sized battery pack).
The link receives its power from an RF carrier sent across the 25 inductive link, and extracts its data from a frequency-modulation (FM) of this carrier. This approach decouples the information from the output voltage, again keeping the output voltage fairly constant for the sake of efficiency. But FM requires a wider bandwidth than that needed for just the data. This link provides that additional 30 bandwidth. This link is still capable of delivering a base-band data rate of one megabit per second. This data rate i~ very high for a neural stimulator. In fact, the band-width limiting element of this system was not the link, but the FM receiver (due to transistors with only a 20 MHz beta cutoff frequency). The stagger-tuned link ~276Z42 can provide both a large bandwidth and a high efficiency; two characteristics which were mutually exclusive in previous links.
Inductive links provide a safe means of delivering both power and data into devices implanted in the body. With the appropriate 5 applications of stagger tùning, these links can now provide a high efficiency by taking advantage of a coupling desensitized link gain.
They can also supply a large bandwidth without adversely effecting their efficiency. This technique is successful in neural stimulators and can easily be applied to many other applications requiring 10 coupling insensitivity.
voltage driver would switch cleanly; and 4 ) to exceed a minimum required voltage gain .
This procedure initially determines whether a design is feasible. If 15 so, then it use6 an iterative process to meet the design goals.
The procedure begins by ~pecifying the type of link, coupling range, load range, and operating frequency. These choices may be limited by size constraints, FCC regulations, or power requirements.
The equation below is an approximation which applies to all of the 20 link combinations. It gives the ratio of the magnitude of the mini-mum gain to the maximum gain (G) in terms of a ratio of the mini-mum and maximum coupling coefficients (Kmin and Kmax). Note that O<K . <K < and O<G<I.
- mm- max- - --G Igainmin I ~ 2~Kmax /Kmin 2~ Igain 1 1+K /K
max max min The coupling range (Kmin and Kmax) is usually found experimental-ly . If the gain variation is too large ( G is too small), then the coupling range needs to be reduced (preferably by raising the min-imum value of coupling which has the added advantages of lowering 30 the transmitter coil losses by reducing the transmitter current, and also improving the load insensitivity of the gain by lowering the link output impedance). One possible solution is to use larger coil ~27~42 diameters. A larger diameter wil] reduce the coupling range as ]ong as the alignment tolerance is held constant. Another fiolution is to use a ferrite backing on either or both coils. The ferrite will compress the magnetic field around the coils and raise their mutual 5 coupling.
Since the gain is still somewhat dependent on coupling, it is neces-sary to find the values of coupling which produce the minimum and maximum values of gain. The minimum gain occurs, by design, at the minimum and maximum values of coupling (K i and ~ ), The 10 maximum gain occurs at the optimum value of coupling given by equation Kopt = KminKmax From here on, the procedure tends to be iterative. Next, the min-imum inductance of each of the two coils is chosen. The receiver coil inductance must provide a reasonable Q (for example, greater 15 than lO) for the loaded receiver tank circuit. If the Q is too low, the output voltage will be sensitive to changes in the load. Also, to maintain a reasonable overall efficiency, the transmitter coi]
inductance should provide a large enough input impedance so that the transmitter will not have to supply an excessive current into the 20 link (generally, raising the transmitter coil's inductance raises the link's input impedance).
The next step is to set the minimum gain and reduce the gain vari-ation over a band of frequencies. A ratio of receiver resonance to 25 the operating frequency (~rl~) is chosen. (Remember, if the link is driven by a class D transmitter, (~r/~) should be chosen so that the input impedance looks inductive, which allows the transmitter to provide efficient switching . ) This ratio along with the resistances, coupling range and coil inductances, fixes the ratio of the transmit-30 ter resonance to the operating frequency (~t/~). The equationswhich specify ~tl~ are different for each type of link and are given below .
~762A2 Next, plots are made of the gain versus frequency for the six com-binations of` minimum and maximum load (Rmjn,RmaX); and minimum, optimum, and maximum values of coupling (kmin,Kopt,lCmaX). The required gain is obtained by varying the turns ratio (inductance of 5 the coils and adjusting ~r/~. For example, if the gain is too small, ~' then the turns ratio should be changed or ~r/~ moved cl4~urc to unity. If the gain variation, due to loading, is too large over the frequency band, then the receiver coil Q should be increased. If the spread is lop-sided (more variation on one side of the band than 10 on the other), then ~r/~ needs to be slightly adjusted until the gain is acceptably uniform about the operating frequency.
There are four combinations of link gain; voltage gain, current gain, transconductance gain, and transimpedance gain. The link gain equation for all four combinations has the form F(K)= AK
-(B K~+C)+j(DK2+E) where k is the coupling coefficient (OCK~ anci A, B, C, D, and E
are variables dependent on the link combination.
J(k)=~ k~ ~/~
where X=B 2~D2 Y=2BC I 2DE
Z=C 2~E2 25 Summarizing for the general section: the minimum gain occurs at k i and kmaX, while the maximum gain occurs at kopt ~mi n~max and is given by f (kopt) ~
The ratio of the minimum gain to the maximum gain is ~27~242 Ikmax 2~
Ge min k max , kmin Therefore, the amount of gain variation (G) and the coupling values where the minimum and maximum gains occur (f in=f (kmin)=f (k ), 10 and fmax=f (kopt) ) are known by simply specifying the coupling range .
Figure 2A is a circuit diagram of a stagger-tuned link which is driven by a voltage source and whose output appears as a voltage source. This arrangement is the type used by the preferred em-15 bodiment. For this circuit the general parameters are ~cl), C~~ Q~, D= ~a 25 E=
u~= 41C~
u~,s= 1 4C, C!, = L~4 ~ ~4 ~276242 Note that ~opt = Z/X = (C2+E2) / (B2+D2) . So substituting andsimplifying give 5 ko~ ZI ~ t~-Solving for the ratio of the operating frequency to the transmitter resonance ~2/~l~t and simplifying gives ~ l-H~V Q~
where H=~
Note the one restriction Qt ~rl Therefore, given the minimum and maximum coupling coefficient, the 20 coil Q's, the operating frequency, and the resonant frequency of the receiver, equation gives the transmitter resonance that minimizes the sensitivity of the gain to the coupling variations. The other three possible stagger-tuned links shown in Fig. 3 follow an equiva-lent approach.
25 Further details of an exemplary embodiment will be given below.
link type: "voltage driven: voltage out"
* coupling range: 0.2~K<0.5 * load range: IKS2'R<3Kn operating frequency: 20 MHz ~ frequency range: 19MHz<f<21MHz transmitter type: class D voltage source * acceptable gain variation across the bandwidth, inc]uding the load variation: 0. 5 drop from the maximum * minimum gain: =2. 0 The ratio of minimum to maximum voltage gain i3 approxima~ely 0 . 90, 5 which is well within the goal of 0. 5 . Next, two identical coi]s, with ferrite backings for high coupling, are chosen. They are 1. 961Jh with a series resistance of 5 . 8Q at 20 MHz . These numbers give a transmitter tank Q of about 40 and a receiver tank Q, including the o l,cn load, of about four.
Next ~ r/w is chosen greater than one so that the input appears inductive for the class D transmitter. Let ~r/~1. 1.
Figure 3B is a p]ot of the gain versus frequency for the six combi-nations of coupling and load, using the equations in the appendix 15 for a "voltage driven:voltage out" link.
Figure 3C is an expanded view of the galn around the operating frequency (20 M~z). The maximurn gain deviation occurs at the high end of the frequency range (Z1 MHz). The ratio of minimum to maximum gain is 0 . 7 which exceeds the design goal of 0. 5 . The 20 minimum gain is about 1.9 which i~ only 5% below the goal of Z, well within the tolerancec of most components, and probably acceptable.
Remember that the receiver Q wa rather low, only 4. The depen-dence of the output voltage on the load is the result of a voltage division between the link output impedance and the effective load 25 impedance. If the link impedance is signi~icant compared to the load, then the output voltage will be load dependent. The curves labeled 1 and 4 on Fi~ure 3C illustrate this point. These curves represent the gain for the minimu~n and maximum load3 at the lowest value o coupling. This link has its highest output irr~pedance when 30 the coils are coupling the least. Over the design band (19 ME1~
21MHz), the large t gain spread occurs at 21 MHz and is bounded by curve~ 1 and 4. Here, the output impedance of the linlc (equiv-alent to about 800 ohms) is significant compared to the load, so the ~L276242 output voltage is somewhat load sensitive. Two ways of reducing the output impedance are: one, raise the coupling coefficient (use ferrite backings and larger coil diameters); and two, make sure that the Q of the loaded receiver coil is reasonably high (adjust the 5 receiver coil inductance).
As mentioned before, this link places some requirements on the driver. Figure 5 shows a preferred embodiment of a transmitter design used to drive the link. This circuit is a class D FM voltage driver with several features for portable, efficient operation. Its 10 frequency is fairly insensitive to changes in the power supply volt-age, so that it can operate from transistor radio batteries without regulation. Its frequency deviation is controlled by a CMOS com-patible input which interfaces easily to portable digital logic. It also uses core drivers in the output stage to rapidly and efficiently 15 switch the high output currents.
The stagger-tuned link, in this example, has a minimum input im-pedance of about 47û, a fairly small load for the transmitter to drive. An efficiency of 35% allows the link to operate for a full day (16 hours) on two nine volt transistor radio batteries. This effici-20 ency provideg a major reduction in power consumption and batterysize over previous Stanford stimulators (a worst case power con-sumption of 140mW instead of 400mW, and two nine volt batteries instead of a brick-sized battery pack).
The link receives its power from an RF carrier sent across the 25 inductive link, and extracts its data from a frequency-modulation (FM) of this carrier. This approach decouples the information from the output voltage, again keeping the output voltage fairly constant for the sake of efficiency. But FM requires a wider bandwidth than that needed for just the data. This link provides that additional 30 bandwidth. This link is still capable of delivering a base-band data rate of one megabit per second. This data rate i~ very high for a neural stimulator. In fact, the band-width limiting element of this system was not the link, but the FM receiver (due to transistors with only a 20 MHz beta cutoff frequency). The stagger-tuned link ~276Z42 can provide both a large bandwidth and a high efficiency; two characteristics which were mutually exclusive in previous links.
Inductive links provide a safe means of delivering both power and data into devices implanted in the body. With the appropriate 5 applications of stagger tùning, these links can now provide a high efficiency by taking advantage of a coupling desensitized link gain.
They can also supply a large bandwidth without adversely effecting their efficiency. This technique is successful in neural stimulators and can easily be applied to many other applications requiring 10 coupling insensitivity.
Claims (14)
1. An inductive link for conveying power and/or data through the skin of a user comprising a first external coil, a second medically implantable coil, the coils being movable relative to one another, the mutual inductive coupling of said coils changing with their position relative to one another, and a means for stagger tuning said first and second coils to maintain a gain which is relatively independent of the coil's mutual coupling at a selected frequency or frequency band of interest.
2. An inductive link as in claim 1 further comprising a class D transmitter means coupled to said first coil for driving said first externally mounted coil.
3. An inductive link as claimed in claim 1 further comprising a voltage regulator coupled across the terminals of said second implanted coil.
4. An inductive link as claimed in claim 1 wherein said coils are circular and of equal size.
5. An inductive link as in claim 4 wherein each of said coils includes a magnet substantially in the center of said coil, the magnets having opposite poles aligned for maintaining alignment of the coils.
6. An inductive data link as in claim 1 wherein each of said means for stagger tuning comprise an RC network, each of said RC networks comprising a filter, the values of each of said RC
network components being selected to tune one of said coils above the frequency of interest, the other of said coils being tuned below the frequency of interest.
network components being selected to tune one of said coils above the frequency of interest, the other of said coils being tuned below the frequency of interest.
7. An inductive link as in claim 6 wherein said means for stagger tuning comprise a series RC network coupled in series with said first input coil and a parallel RC output network coupled in parallel with said second coil to define a voltage in voltage out relationship.
8. An inductive link as in claim 7 further comprising a voltage regulator coupled across the terminals of said second implanted coil.
9. An inductive link as in claim 6 wherein said means for stagger tuning comprise a parallel RC network coupled in parallel with said first coil and a parallel RC network coupled in parallel with said second coil to define a current in, voltage out relationship.
10. An inductive link as in claim 9 further comprising a voltage regulator coupled across the terminals of said second implanted coil.
11. An inductive link as in claim 6 wherein said means for stagger tuning comprise a series RC network coupled in series with said first coil and a series RC network coupled in series with said second coil to define a voltage in: current out relationship.
12. An inductive link as in claim 6 wherein said means for stagger tuning comprise a parallel RC network, coupled in parallel with said first coil and a series RC network coupled in series with said second coil to define a current in, current out relationship.
13. An inductive link as in any one of claims 7 to 12 comprising an FM transmitter means for driving said first coil for transferring power and data to said second coil.
14. An inductive link as in claim 1 including ferrite backing on said first and second coils to maximize coupling between said coils.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US06/719,232 US4679560A (en) | 1985-04-02 | 1985-04-02 | Wide band inductive transdermal power and data link |
| US719,232 | 1985-04-02 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1276242C true CA1276242C (en) | 1990-11-13 |
Family
ID=24889283
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000505519A Expired - Fee Related CA1276242C (en) | 1985-04-02 | 1986-04-01 | Wide band inductive transdermal power and data link |
Country Status (7)
| Country | Link |
|---|---|
| US (1) | US4679560A (en) |
| EP (1) | EP0200359B1 (en) |
| JP (1) | JPS625358A (en) |
| CN (1) | CN1005456B (en) |
| AU (2) | AU5535186A (en) |
| CA (1) | CA1276242C (en) |
| DE (1) | DE3686194D1 (en) |
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| US3195540A (en) * | 1963-03-29 | 1965-07-20 | Louis C Waller | Power supply for body implanted instruments |
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| JPS5060085A (en) * | 1973-09-27 | 1975-05-23 | ||
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| US4134408A (en) * | 1976-11-12 | 1979-01-16 | Research Corporation | Cardiac pacer energy conservation system |
| US4441210A (en) * | 1981-09-18 | 1984-04-03 | Hochmair Erwin S | Transcutaneous signal transmission system and methods |
-
1985
- 1985-04-02 US US06/719,232 patent/US4679560A/en not_active Expired - Fee Related
-
1986
- 1986-03-26 DE DE8686302272T patent/DE3686194D1/en not_active Expired - Lifetime
- 1986-03-26 EP EP86302272A patent/EP0200359B1/en not_active Expired - Lifetime
- 1986-03-27 AU AU55351/86A patent/AU5535186A/en not_active Abandoned
- 1986-03-31 CN CN86103021.4A patent/CN1005456B/en not_active Expired
- 1986-04-01 CA CA000505519A patent/CA1276242C/en not_active Expired - Fee Related
- 1986-04-02 JP JP61076406A patent/JPS625358A/en active Pending
-
1990
- 1990-05-23 AU AU55788/90A patent/AU5578890A/en not_active Abandoned
Also Published As
| Publication number | Publication date |
|---|---|
| EP0200359A3 (en) | 1987-12-02 |
| CN86103021A (en) | 1986-11-12 |
| DE3686194D1 (en) | 1992-09-03 |
| EP0200359B1 (en) | 1992-07-29 |
| AU5578890A (en) | 1990-09-20 |
| CN1005456B (en) | 1989-10-18 |
| US4679560A (en) | 1987-07-14 |
| JPS625358A (en) | 1987-01-12 |
| EP0200359A2 (en) | 1986-11-05 |
| AU5535186A (en) | 1986-10-09 |
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