CA1188740A - Vhf sensor in-band radio relay - Google Patents
Vhf sensor in-band radio relayInfo
- Publication number
- CA1188740A CA1188740A CA000416626A CA416626A CA1188740A CA 1188740 A CA1188740 A CA 1188740A CA 000416626 A CA000416626 A CA 000416626A CA 416626 A CA416626 A CA 416626A CA 1188740 A CA1188740 A CA 1188740A
- Authority
- CA
- Canada
- Prior art keywords
- signal
- signals
- circuit means
- coupled
- radio apparatus
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
Links
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B7/00—Radio transmission systems, i.e. using radiation field
- H04B7/14—Relay systems
- H04B7/15—Active relay systems
- H04B7/155—Ground-based stations
- H04B7/15564—Relay station antennae loop interference reduction
- H04B7/15585—Relay station antennae loop interference reduction by interference cancellation
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B7/00—Radio transmission systems, i.e. using radiation field
- H04B7/14—Relay systems
- H04B7/15—Active relay systems
- H04B7/155—Ground-based stations
- H04B7/15528—Control of operation parameters of a relay station to exploit the physical medium
- H04B7/1555—Selecting relay station antenna mode, e.g. selecting omnidirectional -, directional beams, selecting polarizations
Abstract
ABSTRACT
Disclosed is an in-band, non-sampling, real time VHF radio relay or signal repeater for use in two-way, multi hop remote sensor data links.
Each signal repeater comprises a transceiver which includes interference cancellation circuits for preventing the high level signal produced by the radio repeater;s transmitter section from desensitizing the low noise RF
amplifier circuitry incorporated in the front end of the radio repeater's receiver section. Such apparatus is utilized to implement a remote sensor data collection network consisting of a plurality of subject radio repeaters and several sensors normally arranged in groups. The data which flows over the relay network can either originate at the sensors and flow to a read out station or it may originate at the read out station and flow to the sensors.
Sensor originated information may be either digital or analog, or both, while read-out station originated information is digital only. The relay operates with either analog or digitally-modulated signals. When used with an omni-directional antenna, it is not direction-preferential.
Disclosed is an in-band, non-sampling, real time VHF radio relay or signal repeater for use in two-way, multi hop remote sensor data links.
Each signal repeater comprises a transceiver which includes interference cancellation circuits for preventing the high level signal produced by the radio repeater;s transmitter section from desensitizing the low noise RF
amplifier circuitry incorporated in the front end of the radio repeater's receiver section. Such apparatus is utilized to implement a remote sensor data collection network consisting of a plurality of subject radio repeaters and several sensors normally arranged in groups. The data which flows over the relay network can either originate at the sensors and flow to a read out station or it may originate at the read out station and flow to the sensors.
Sensor originated information may be either digital or analog, or both, while read-out station originated information is digital only. The relay operates with either analog or digitally-modulated signals. When used with an omni-directional antenna, it is not direction-preferential.
Description
8~
13ACKGROUND OI: T~IE INV~NT[O~
_ _ . _ _ rhis investion relates generally to radio transceiver apparatus and more particularly to :in-band, radio relay type transceiver apparatus.
In the dep:Loyment o:F a remote sensor-data collection system it i9 frequently necessary to include a number of raclio repecIter <3ites to overcorne the line of sight transmission restrictions, thereby e.~tending the range of data collection. The :Eundamental probl.em, however, i.n implementi.ng flny radio relay is in the pre~erItillg of the relativel.y high power signa~ :Erom the re1.ay~s transmitter from being picked up by i.ts own receiver at a l.evel h:igII enough to desen~q:itize it. Pr:Lor art raclio relays have utiliæed a vari.ety oE tecII-niques to achieve the required transmitter to receiver isolat:ion, :Eor example, a conventional out-of-band re:Lay is implementecl with the transmitter frequeIlcy ft and receiver frequency f being in di.fferent bands and with isolation being achieved by means of fixed ft and f band filte:r.s. The problems associated with such an apparatus, however, is -the frequency allocation problem which presently has many undesirable restrictions. With respect to the conventional in-band relay system, a relatively large ft and f frequency separation exists between the operational transmi.t and receive frequencies.
However in such apparatus t~mable narrow band filters are utilized which become relatively large and expensive and many frequency channels are inherently unusab]e. Another type of radio relay system known to those skilled in the art is the store and forward relay type of system which alternately receives and transmits each message. The problem associated with this type of system is that the relays require a data storage capability which encounters severe problems where analog data is being stored and also only a 50% receiver duty cycle is possible~ Another known type of rel.ay system comprises what is known as a sampling relay wherein the message is alternately received and transmitted at a Nyquist sampling rate. Such a system has an inherent limitation that it is subject to spectrum splatter and loss o.E isolation in ground environments due to delayed reflections. All of the above noted radio relays typically ; include a common antenna which is shared by both the transmitter and receiver ., portions oE the relcsy. Also known are radio relay systems wllich operate at clny transrnit and receive frecl~lency but these systems inclu(le respective directional antennas whicll are highly directive. In such systerns no omni-directional coverage is obtainable clnd the sntenna3 tllemselve6 tend to become relatively large at VHI~ frequencies.
Accorclingly, it is an object oE the l~rescllt invenl:ion to provide an improvement in radio reLay apparrltus.
It is anottler object of the present illVellt:iOII to provide an in-bflnd radio relay Apparfltus which incLucles improved iso]cltion b~tween transrnitter flnd receiver.
It is yet another object of the present invention Lo provide an in-band radio relay apparatus which includes adaptive coherent interference cancellation to prevent the transmitter from desensitizing the receiver during operation.
~ hese and other objects are accomplished by means of an in-band, non-sampling, real time radio relay or signal repeater apparatus for use in two-way multi-hop remote sensor data links, each signal repeater apparatus having at least one interference canceliation circuit or sub-system in the receiver section which is operable in response to its own transrnitted RF
reference signal and the received RF signal appearing at a common receive/
transmit antenna, whereupon the RF reference signal's phase and amplitude are adaptively adjusted and summed with the received RF signal to effect can-cellation of most of the transmitted RF signal appearing at the -front end of the receiver. The received RF signal is down-converted to an IF signal for narrowband filtering, and then it is up-converted to an offset transmit frequency, amplified to a fixed level, snd fed back to the common antenna through a bandpass filter and switched attenuator to provide the RF trans-mitted signal and from which the RF reference signal is derived. When desirable, further cancellation of the transmitted signal cancellation residue is further provided prior to the RF signal down-conversion. Additionally, t~
DC power i8 conserved by inclu(l:ing an RF signal preserlce detector Means in the receiver port:ion wh:icll operates to enabl.e the DC power supply which is used to power the transmi.tte portion upon RF signal presence being detected.
DESCRIPTION OF T~IE DRAWINGS
Figure l i9 a block di.agram generally i1.1~lstrative of a data collection network utilizing the present invention;
Figure 2 i8 an electrical block diagram generally i.l.lustrative of the preferred embodiment of the subj(lct :I.nventi.on;
Figure 3 :Ls an electr:ical b:Lock diagrarn generally i].lustrative of an interfererlce cancellat:lon circuit. inclucled in the embodiment o:E the invention shown in Flgure 2;
Figure ~ is an electrical bJ.ock diagram of the complex weight circuitry shown in Figure 3;
Figure 5 is an electrical schematic diagram of tlle bi-polar attenuators included in the complex weight circuitry shown in Figure 4;
Figure 6 is an electrical schematic diagram of the driver circuitry utilized for the bi-polar attenuators shown in Figure 5; and Figure 7 is an electrical schematic diagram of the co~plex weight control unit included in the interference cancellation circuitry shown in Figure 3.
DESCRIPTION OF THE PRF.FERRED BUBODI~I~NTB
Referring now to the drawing~s wherein like reference numerals refer to like components throughout, attention is first directed to Figure 1 wherein there is shown diagrammatically a typical remote sensor data collection net-work comprised of four sensor in-band radio re].ay (SIRR) units 101, 102, 103, 104 operating with respective sensor groups 121, 122, 123, and 124 each including a plurality of sensors 14. At one end of the network is a read-out unit 16. Information which :Elows through the repeater network consisting of ; the units 101 .... 10~ originates at any of the sensor 14 and flows to the read-out unit 16 via one or more re].ay units or it may originate at the read-out un~t 1.6 v:i.a ont or more re].ay units or it rnay or;.ginaLe at the read-out un:i,t 16 and fl.ow to the serlsors 14. Sensor or:i.g:inat,tecl in-formation i8 commonly re~erred to as "seni:or data" and may b~ e:itl~er ~igitaL or ana:log in nature, o-r both, while read out uni.t or:ig:i.nated irlformation i9 COmmOtlly referred to as a "sensor commalld" and :is only di,gitcl :in nature~
Each SIRR unit 10i is designed to rel.ay a s:igna], i.n the V~IF range, offsetting the frequency oL retransnlissi.on ft from the frequency Or reception f by a predetermined amount, i.e. [t ~l~f- 'ryp:i.clll:l.y f is :Ln the range be-tween 160Mllz and 176MI-lz whi]e ~f is as mucll as 1.6M}Iz or as ll.ttlc a~; 93k~1z.
F.ach SIRR ~Init lOJ relays i.n real time analog or d:Lg:ital FM slgnals with a bandwidth of up to 18k~1z. To obtain higll :Eorward gai.n, however, isolation between the transmitter and receiver porti.ons is necessary in order to obtai.n a stable operation. 'l'his is achieved by the use of an adaptive coherent inter-ference cancellation sub-system in combination with narrow band fi]tering and antenna isolation as will be shown as the present detailed description con-tinues.
Referring now to Figure 2, shown therein is a simplified block diagram generally illustrative of one unit 10 of the sensor in-band radio relay (SIRR) UllitS 101 ... 10~ shown in Figure 1. The SIRR unit 10 is com-prised of a receiver section 18 and a transmitter section 2() coupled to a common transmit/receive antenna 22. Considering fi.rst the receiver section 18, a first interference cancellati.on circui.t or sub-system 24 is provided in : ~ order to prevent the high level RF signal currently being retransmitted by the transmitter section 20 from desensitizing a relatively low noise RF ampli-fier 26 to the RF signal concurrently being received while a second inter-ference cancellation sub-system 28 is included -to provide a further cancella-tion of any transmitter signal residue not eliminated by the first inter-ference cancellation sub-system 24. Both interference cancellation sub-system : ~ 24 and 28 are substantially indentical in construction and are shown in detail in Figures 3 through 8.
As shown in F:igure 2, however, the f;.rst interference cancellation sub-system 24 is coupled to the antemla 22 through a signal coupler 30.
Accordingly the received RF si.gnal along with any leakage of the signal being transmitted as well as any reLlections from surrounding terrain is also coupled through the coupler 30. A second signal coupler 32 is utilized and is adapted to provide a -transmitted RF re:ference signal which is also applied to the interference cancellation sub-system 24. As will be shown, the phase and amplitude of this RF reference signal is adaptivel.y adjusted in the interference cancellation sub-system 24 to effect cancellation of the undesired retransmittecl signal coming through the coupler 30 without any cancellation of the receive signal appeari.ng thereat. The uncancelled received RF signal plus the residual retransmitted signal are then applied to the amplifier 26 wherein RF arnplification takes place. The output o-f the RF amplifier 26 is fed to the second interference cancellation sub-system 28 along with the transmitted signal reference appearing at the coupler 32 and conveyed via the transmission line 34~ As noted earlier, the operation of the second interference cancellation sub-system 28 is to further reduce any residue of the transmitted signal not fully cancelled in the suh-system 24.
The output of the interference cancellation sub-system 28 is fed to a signal mixer 36 along with a local oscillator signal generated by a local. oscillator 38 whereupon the received RF signal is down-converted to an IF signal where it is fed to a crystal bandpass filter 40 which is adapted to provide channel selectivity with rejection of any out-of-channel signals. The filtered IF
signal is next fed to an IF amplifier 42 which is operable to raise the gain of the IF signal to the level sufficient to energize a received ~F signal presence detector 44 which i.s adapted to enable a DC power supply 46 which powers the transmitter section 20 so that prior to any received signal being detected no DC power is applied to the transmitter section 20 in order to conserve power.
After rece:ived RF signal presence detection occurs, the IF signa].
which i simultaneously applied to a second IF amplifier locatecl in the trans-mitter section 20 I.s ampl.if-ied and app:L:ied to another crystal bandpass filter 50 in order to conf:ine the system noise to the retransmitted si.gnal channel.
The IF s:ignal thus filtered is then up-converted to a selected transmit frequency Et by being appli.ed to a signa1. mixer 52 along with a transmitter local osc:il.lator signal generated by the local oscil]ator 54. The up conversion local oscillator frequency :is selected to oEfset the R~` signal outputted from the mixer 52 by a predetermined increment~ f from the origirlally received signal frequency f . The output oE the l-nixer 52 i9 red to an RF
ampl.ifier 55 and then to a bandpass fi.lter 56 where it is coupl.ed back to the common transmit/receive antenna 22 through a switched attenuator 58 whi.ch is adapted to provide a high power or low powe-r operationa:L mode. Thus each SIRR unit 101 ... L0~ is adapted to receive an incoming RF si.gna]. and then suitably convert the signal to an o:Efset E~F frequency and retransmi.t i.t over the network (Figure 1~ with the receiver section 18 operating at a 100% duty cycle without being swamped by the signal which is being fed back through the antenna 22 from the transmitter section 20, ; At the heart of the successful operation of each of the SIRR units 101 ... 104 i5 the use of a-t least one interference cancellation sub-system (ICS) 24 to protect the receiver section lo from interference by the co-located transmitter section 20. The apparatus employed by the subject invention is broadly shown by the block diagram of F:igure 3. Because of imperfect isolation in the antenna network 60 which includes the signal coupler 30 shown in Figure
13ACKGROUND OI: T~IE INV~NT[O~
_ _ . _ _ rhis investion relates generally to radio transceiver apparatus and more particularly to :in-band, radio relay type transceiver apparatus.
In the dep:Loyment o:F a remote sensor-data collection system it i9 frequently necessary to include a number of raclio repecIter <3ites to overcorne the line of sight transmission restrictions, thereby e.~tending the range of data collection. The :Eundamental probl.em, however, i.n implementi.ng flny radio relay is in the pre~erItillg of the relativel.y high power signa~ :Erom the re1.ay~s transmitter from being picked up by i.ts own receiver at a l.evel h:igII enough to desen~q:itize it. Pr:Lor art raclio relays have utiliæed a vari.ety oE tecII-niques to achieve the required transmitter to receiver isolat:ion, :Eor example, a conventional out-of-band re:Lay is implementecl with the transmitter frequeIlcy ft and receiver frequency f being in di.fferent bands and with isolation being achieved by means of fixed ft and f band filte:r.s. The problems associated with such an apparatus, however, is -the frequency allocation problem which presently has many undesirable restrictions. With respect to the conventional in-band relay system, a relatively large ft and f frequency separation exists between the operational transmi.t and receive frequencies.
However in such apparatus t~mable narrow band filters are utilized which become relatively large and expensive and many frequency channels are inherently unusab]e. Another type of radio relay system known to those skilled in the art is the store and forward relay type of system which alternately receives and transmits each message. The problem associated with this type of system is that the relays require a data storage capability which encounters severe problems where analog data is being stored and also only a 50% receiver duty cycle is possible~ Another known type of rel.ay system comprises what is known as a sampling relay wherein the message is alternately received and transmitted at a Nyquist sampling rate. Such a system has an inherent limitation that it is subject to spectrum splatter and loss o.E isolation in ground environments due to delayed reflections. All of the above noted radio relays typically ; include a common antenna which is shared by both the transmitter and receiver ., portions oE the relcsy. Also known are radio relay systems wllich operate at clny transrnit and receive frecl~lency but these systems inclu(le respective directional antennas whicll are highly directive. In such systerns no omni-directional coverage is obtainable clnd the sntenna3 tllemselve6 tend to become relatively large at VHI~ frequencies.
Accorclingly, it is an object oE the l~rescllt invenl:ion to provide an improvement in radio reLay apparrltus.
It is anottler object of the present illVellt:iOII to provide an in-bflnd radio relay Apparfltus which incLucles improved iso]cltion b~tween transrnitter flnd receiver.
It is yet another object of the present invention Lo provide an in-band radio relay apparatus which includes adaptive coherent interference cancellation to prevent the transmitter from desensitizing the receiver during operation.
~ hese and other objects are accomplished by means of an in-band, non-sampling, real time radio relay or signal repeater apparatus for use in two-way multi-hop remote sensor data links, each signal repeater apparatus having at least one interference canceliation circuit or sub-system in the receiver section which is operable in response to its own transrnitted RF
reference signal and the received RF signal appearing at a common receive/
transmit antenna, whereupon the RF reference signal's phase and amplitude are adaptively adjusted and summed with the received RF signal to effect can-cellation of most of the transmitted RF signal appearing at the -front end of the receiver. The received RF signal is down-converted to an IF signal for narrowband filtering, and then it is up-converted to an offset transmit frequency, amplified to a fixed level, snd fed back to the common antenna through a bandpass filter and switched attenuator to provide the RF trans-mitted signal and from which the RF reference signal is derived. When desirable, further cancellation of the transmitted signal cancellation residue is further provided prior to the RF signal down-conversion. Additionally, t~
DC power i8 conserved by inclu(l:ing an RF signal preserlce detector Means in the receiver port:ion wh:icll operates to enabl.e the DC power supply which is used to power the transmi.tte portion upon RF signal presence being detected.
DESCRIPTION OF T~IE DRAWINGS
Figure l i9 a block di.agram generally i1.1~lstrative of a data collection network utilizing the present invention;
Figure 2 i8 an electrical block diagram generally i.l.lustrative of the preferred embodiment of the subj(lct :I.nventi.on;
Figure 3 :Ls an electr:ical b:Lock diagrarn generally i].lustrative of an interfererlce cancellat:lon circuit. inclucled in the embodiment o:E the invention shown in Flgure 2;
Figure ~ is an electrical bJ.ock diagram of the complex weight circuitry shown in Figure 3;
Figure 5 is an electrical schematic diagram of tlle bi-polar attenuators included in the complex weight circuitry shown in Figure 4;
Figure 6 is an electrical schematic diagram of the driver circuitry utilized for the bi-polar attenuators shown in Figure 5; and Figure 7 is an electrical schematic diagram of the co~plex weight control unit included in the interference cancellation circuitry shown in Figure 3.
DESCRIPTION OF THE PRF.FERRED BUBODI~I~NTB
Referring now to the drawing~s wherein like reference numerals refer to like components throughout, attention is first directed to Figure 1 wherein there is shown diagrammatically a typical remote sensor data collection net-work comprised of four sensor in-band radio re].ay (SIRR) units 101, 102, 103, 104 operating with respective sensor groups 121, 122, 123, and 124 each including a plurality of sensors 14. At one end of the network is a read-out unit 16. Information which :Elows through the repeater network consisting of ; the units 101 .... 10~ originates at any of the sensor 14 and flows to the read-out unit 16 via one or more re].ay units or it may originate at the read-out un~t 1.6 v:i.a ont or more re].ay units or it rnay or;.ginaLe at the read-out un:i,t 16 and fl.ow to the serlsors 14. Sensor or:i.g:inat,tecl in-formation i8 commonly re~erred to as "seni:or data" and may b~ e:itl~er ~igitaL or ana:log in nature, o-r both, while read out uni.t or:ig:i.nated irlformation i9 COmmOtlly referred to as a "sensor commalld" and :is only di,gitcl :in nature~
Each SIRR unit 10i is designed to rel.ay a s:igna], i.n the V~IF range, offsetting the frequency oL retransnlissi.on ft from the frequency Or reception f by a predetermined amount, i.e. [t ~l~f- 'ryp:i.clll:l.y f is :Ln the range be-tween 160Mllz and 176MI-lz whi]e ~f is as mucll as 1.6M}Iz or as ll.ttlc a~; 93k~1z.
F.ach SIRR ~Init lOJ relays i.n real time analog or d:Lg:ital FM slgnals with a bandwidth of up to 18k~1z. To obtain higll :Eorward gai.n, however, isolation between the transmitter and receiver porti.ons is necessary in order to obtai.n a stable operation. 'l'his is achieved by the use of an adaptive coherent inter-ference cancellation sub-system in combination with narrow band fi]tering and antenna isolation as will be shown as the present detailed description con-tinues.
Referring now to Figure 2, shown therein is a simplified block diagram generally illustrative of one unit 10 of the sensor in-band radio relay (SIRR) UllitS 101 ... 10~ shown in Figure 1. The SIRR unit 10 is com-prised of a receiver section 18 and a transmitter section 2() coupled to a common transmit/receive antenna 22. Considering fi.rst the receiver section 18, a first interference cancellati.on circui.t or sub-system 24 is provided in : ~ order to prevent the high level RF signal currently being retransmitted by the transmitter section 20 from desensitizing a relatively low noise RF ampli-fier 26 to the RF signal concurrently being received while a second inter-ference cancellation sub-system 28 is included -to provide a further cancella-tion of any transmitter signal residue not eliminated by the first inter-ference cancellation sub-system 24. Both interference cancellation sub-system : ~ 24 and 28 are substantially indentical in construction and are shown in detail in Figures 3 through 8.
As shown in F:igure 2, however, the f;.rst interference cancellation sub-system 24 is coupled to the antemla 22 through a signal coupler 30.
Accordingly the received RF si.gnal along with any leakage of the signal being transmitted as well as any reLlections from surrounding terrain is also coupled through the coupler 30. A second signal coupler 32 is utilized and is adapted to provide a -transmitted RF re:ference signal which is also applied to the interference cancellation sub-system 24. As will be shown, the phase and amplitude of this RF reference signal is adaptivel.y adjusted in the interference cancellation sub-system 24 to effect cancellation of the undesired retransmittecl signal coming through the coupler 30 without any cancellation of the receive signal appeari.ng thereat. The uncancelled received RF signal plus the residual retransmitted signal are then applied to the amplifier 26 wherein RF arnplification takes place. The output o-f the RF amplifier 26 is fed to the second interference cancellation sub-system 28 along with the transmitted signal reference appearing at the coupler 32 and conveyed via the transmission line 34~ As noted earlier, the operation of the second interference cancellation sub-system 28 is to further reduce any residue of the transmitted signal not fully cancelled in the suh-system 24.
The output of the interference cancellation sub-system 28 is fed to a signal mixer 36 along with a local oscillator signal generated by a local. oscillator 38 whereupon the received RF signal is down-converted to an IF signal where it is fed to a crystal bandpass filter 40 which is adapted to provide channel selectivity with rejection of any out-of-channel signals. The filtered IF
signal is next fed to an IF amplifier 42 which is operable to raise the gain of the IF signal to the level sufficient to energize a received ~F signal presence detector 44 which i.s adapted to enable a DC power supply 46 which powers the transmitter section 20 so that prior to any received signal being detected no DC power is applied to the transmitter section 20 in order to conserve power.
After rece:ived RF signal presence detection occurs, the IF signa].
which i simultaneously applied to a second IF amplifier locatecl in the trans-mitter section 20 I.s ampl.if-ied and app:L:ied to another crystal bandpass filter 50 in order to conf:ine the system noise to the retransmitted si.gnal channel.
The IF s:ignal thus filtered is then up-converted to a selected transmit frequency Et by being appli.ed to a signa1. mixer 52 along with a transmitter local osc:il.lator signal generated by the local oscil]ator 54. The up conversion local oscillator frequency :is selected to oEfset the R~` signal outputted from the mixer 52 by a predetermined increment~ f from the origirlally received signal frequency f . The output oE the l-nixer 52 i9 red to an RF
ampl.ifier 55 and then to a bandpass fi.lter 56 where it is coupl.ed back to the common transmit/receive antenna 22 through a switched attenuator 58 whi.ch is adapted to provide a high power or low powe-r operationa:L mode. Thus each SIRR unit 101 ... L0~ is adapted to receive an incoming RF si.gna]. and then suitably convert the signal to an o:Efset E~F frequency and retransmi.t i.t over the network (Figure 1~ with the receiver section 18 operating at a 100% duty cycle without being swamped by the signal which is being fed back through the antenna 22 from the transmitter section 20, ; At the heart of the successful operation of each of the SIRR units 101 ... 104 i5 the use of a-t least one interference cancellation sub-system (ICS) 24 to protect the receiver section lo from interference by the co-located transmitter section 20. The apparatus employed by the subject invention is broadly shown by the block diagram of F:igure 3. Because of imperfect isolation in the antenna network 60 which includes the signal coupler 30 shown in Figure
2, the voltage standing wave ratio (VSWR~ of the antenna 22 and reflections from nearby objects, a small portion T of the retransmitted signal is coupled back into the interference cancellation sub-system 24 located in the receiver via transmission line 620 The power amplitude of the signal T at this point is relatively larger than the received signal R as shown by the graphical ; illustration 94~ Also shown in Figure 3 is a portion of the retransmitted signal which is cou-pled to ICS 24 by means of the signal coupler 32. This signal is utilized as an RF reference signal and designated REF. The ICS 24 includes a complex weight circuit 66, a complex weight control unit 68, a signal summer 70 and a directional coupler 72. A weighted reference signal REF' which comprises the si.gnal REF suitably altered i.n phase and amplitude is outputted from the complex weight circuit 66 where i.t is algebraically combined with the composite RF signal appearing on line 62 to provide an error signal on line 71 which is fed back to the weight control unit 68. The unit 68 effects the proper setti.ng of the complex weight circuit 66 to allow the signal T to be cancelled while the receive signal R is maintained at the original level as indicated by the graphical illustration 74. The control unit 68 in actuality comprises a complex correlator which is adapted to correlate the error signal from the coupler 72 with the reference signal from the coupler 32 and will be explained when Figure 7 is considered. Thus the summation circuit 70 is provided with a replica o:E the retransmitted signal whose amplitude and phase have been adjusted in the complex weight circuit 66 to effect cancellation of the T signal at its output. Furthermore, attenuation via filtering is provided in the compl.ex weight circuit 66 so that noise side-bands placed on the retransmitted signal by circuit noise modulation are reduced so that they also do not desensitize the recelver section 18.
P~eferring now to the details of the interference cancellation sub-system 24, reference is now made to Figure 4 where the block diagram of the complex weighting circuit 66 is shown including a quadrature hybrid signal coupler 78 which accepts the RF reference signal REF applied to input terminal ôO and splits the signal into inphase I and quadrature Q component signals which appear on transmission lines 82 and 84. The I and Q reference signals are coupled to respective bi-polar attenuators 86 and 88 whose details are shown in Figure 5. The bi polar attenuators 86 and 88 are controlled in accordance with the operation of respective I and Q driver circuits 90 and 92, whose details are shown in Figure 6. The bi-polar attenuators 86 and 88 operate to adjust their respective inputs in amplitude with either positive or negative polarity and provide output signals on signal lines 94 and 96 where they are then combined in phase in a signal combiner 98 to provide the signal REF~ at output terminal 100 which provides an RF signal corresponding to the reference signal REF which is altered in phase and a~lpLi~:ude and applied to the signal summer 70 shown in Figure 3. In addition to the elements noted, the complex weight circuit shown in Figure 4 also includes signal couplers 102 and 104 in the I and Q signal ]ines 82 and 84 ahead of the attenuators 86 and 88 to provide a portion of respective I and Q quad-rature signals to terminals 106 and 108 which are adapted to be coupled to the complex correlator shown in Figure 7 at the input terminals 110 and 112, respectively.
Schematically, the bi-polar attenuators 86 and 88 are identical with attenuator 86 being shown in Figure 5. The circuitry includes four PIN
diodes CRl, CR2, CR3 and CR4 arranged in a bridge configuration between input and output transformers 114 and 116. RF coupling is achieved by means of the capacitors 118, 120, 122, 124, 126 and 128. Bias current is applied to the diodes CRl and CR2 by means of a driver current I applied to terminal "a"
by means of inductances 130 and 132. In an identical fashion, bias current is applied to CR3 and CR4 by means of a driver current Ib applied to terminal "b" via the inductances 134 and 136. The circuit shown in Figure 5 is operable such that when CRl and CR2 are biased to have relatively low RF resistance, the diodes CR3 and CR4 are biased to have relatively high RF resistance and the bi-polar attenuator provides minimum attenuation at output terminal 138 without signal inversion. When diodes CRl and CR2, on the other hand, are biased to exhibit relatively high RF resistance, diodes CR3 and CR4 are biased to exhibit a relatively low RF resistance, and the bi-po]ar network provides minimum attenuation with signal inversion. ~hen all four diodes CRl, CR2, CR3 and CR4 are biased to have equa] RF resistance, the bridge is balanced and maximum attenuation is provided at terminal 138 which couples to either signal line 94 or 96 shown in Figure 4.
The terminal "a" and "b" of the bi-polar attenuators are connected to respective driver circuits 90 and 92, one of which is shown in Figure 6. As shown in Eigure 6, the driver circuit 90 includes an I driver control voltage input terminal 140 which receives a control signal from the I driver control output terminal 142 of the complex correlator shown in Figure 7. The driver ~L~81~7~
circuit 90 further inclucles a first pair of transistors Ql and Q2 coupled to a resistor-diode network 142 including diodes CR5, CR6, and CR7 and a second pair of transistors Q3 and Q4 coupled to a second resistor-diode network 144 including diodes CR8, CR9 and CR10. Transistors Ql and Q3 operate as emitter-follower transistors with the drive currents I and Ib emanating from the collectors of transistors Q2 and Q4, respectively. An operational amplifier 145 is inserted between the input terminal 140 and the emitter-follower Q3 in order to provide a unity again inverting amplifier and thus provide currents I and Ib whose variations are of mutually opposite polarity.
The resistor-diode networks 142 and 144 connected in the emitters of transistors Q2 and Q4 provide non-linear shaping of the control currents I
and Ib, respectively, to compensate for the non-linear control characteristics of the PIN diodes CRl ... CR4 of the bi-polar attenuator 86. The level of the I driver control input voltage applied to the input termina:L 140 controls the output currents of the drive transistors Q2 and Q~ in order to vary the RF
attenuation exhibited by the bi-polar attenuator 86. It should be noted, how-ever, that the resistor-diode networks 142 and 144 each include a variable resistor 146 and 148, respectively, which are used to set the current values of I and Ib for maximum RF attenuation when the amplitude of the con-trol voltage applied to terminal 140 is substantially mid-way in its input range, for example, if the input voltage varies from 1 to 5 volts, the resistors 146 and 148 are adjusted to provide for tnaximum attenuation when the control voltage is in the region of 2.5 to 3 volts. Additionally, transistors Q2 and Q4 are powered from a t 6V supply which is turned on only when the transmitter section 20 (Figure 2) is enabled. The circuit configurations for the bi-polar attenuator 88 and the driver 92 are identical and their operation is the same as described above and therefore need not be repeated.
Referring now to Figure 7, the I and Q driver control voltages for the complex weight circuit 66 shown in Figure 3 are derived by a weight control unit 68 which consists of a circuit which implements a complex correlation between the I and Q quadrature components of the reference signal REF taken ~From the coupler 102 and 104 (Figure 4), and the error signal taken from the coupler 72 shown in Figure 3. Accordingly, as shown in Figure 7, the in-phase I and quadrature Q components oE the reference signal REF from the complex weighting circuit are appliecl to input terminals 110 and 112 while the error signal from the coupler 72 is applied to input terminal 150. The error signal input is cormected to a power divider 152 which provides output signal lines 154 and 156 which respectively are connected to a pair of signal mixers 158 and 160 along with I and Q components of the reference signal REF. As shown, mixer 158 is adapted to correlate the I component while the mixer 160 is adapted to correlate the Q component. In addition, a DC bias is applied to the mixers 158 and 160 at termina]s 162 and 164, respectively, so that the out-put therefrom which appears on signal leads 166 and 168 rides on a DC :Level corresponding to the amplitude of the bias vo]tage applied to the mixers. The output of the I component mixer 158 is fed to an operational amplifier 170 which includes a low pass filter network 172 coupled in a feedback arrangement around the amplifier. Accordingly, the driver control signal for the in-phase I driver ciruit 90 shown in Figure 4 is coupled from output terminal. 142 to the I driver control input terminal 140 of Figure 4. Additionally, operational amplifier 170 includes a grounded variable resistance element 174 which pro~
vides a DC offset adjustment for the level of the output control voltage appearing at terminal 142. In a like manner, the quadrature Q component of the reference signal ~EF which is correlated in the mixer 160 is applied to an operational amplifier 176 having the filter feedback network 178 coupled thereto to provide a filtered amplified driver signal to the Q component driver 92 shown in Figure 4 by a suitable connection of terminal l44 and as shown in Figure 7 to terminal 141 shown in Figure 4. DC offset adjustment of the output from the amplifier 176 is provided by means of the grounded variable res:istance element 180. Also both operational amplifiers 170 and 176 are operated as differential amplifiers which have their respective + inputs connected to a reference voltage of ~3V applied t terminals 182 and 184, respectively, and are adapted to drive the output signal amplitude and phase . . .
7f~
from comple~ weighting ci.rcu:it 66 to the values needed to cancel the retrans-mitted signal.
Thus what has been shown and described is an i.n-band non-sampling real-time VHF radio repeater/relay tra[lscei.ver for use in two-way multi-hop remote sensor clata links whlch includes unique interference cancellation circuitry which operates to prevent the high level transmitted signal produced by the transmitter section of the transceiver from desensitizing the low noise RF amplifier located in the :Front of the receiver sectlon of the transceiver.
: While the invention has been particularly shown and described with reference to a preferred embodiment thereof, it is to be understood from the foregoing that other changes and modifications in the form and details may be made without departing from the spirit and scope of the invention as set forth in the subtended claims.
P~eferring now to the details of the interference cancellation sub-system 24, reference is now made to Figure 4 where the block diagram of the complex weighting circuit 66 is shown including a quadrature hybrid signal coupler 78 which accepts the RF reference signal REF applied to input terminal ôO and splits the signal into inphase I and quadrature Q component signals which appear on transmission lines 82 and 84. The I and Q reference signals are coupled to respective bi-polar attenuators 86 and 88 whose details are shown in Figure 5. The bi polar attenuators 86 and 88 are controlled in accordance with the operation of respective I and Q driver circuits 90 and 92, whose details are shown in Figure 6. The bi-polar attenuators 86 and 88 operate to adjust their respective inputs in amplitude with either positive or negative polarity and provide output signals on signal lines 94 and 96 where they are then combined in phase in a signal combiner 98 to provide the signal REF~ at output terminal 100 which provides an RF signal corresponding to the reference signal REF which is altered in phase and a~lpLi~:ude and applied to the signal summer 70 shown in Figure 3. In addition to the elements noted, the complex weight circuit shown in Figure 4 also includes signal couplers 102 and 104 in the I and Q signal ]ines 82 and 84 ahead of the attenuators 86 and 88 to provide a portion of respective I and Q quad-rature signals to terminals 106 and 108 which are adapted to be coupled to the complex correlator shown in Figure 7 at the input terminals 110 and 112, respectively.
Schematically, the bi-polar attenuators 86 and 88 are identical with attenuator 86 being shown in Figure 5. The circuitry includes four PIN
diodes CRl, CR2, CR3 and CR4 arranged in a bridge configuration between input and output transformers 114 and 116. RF coupling is achieved by means of the capacitors 118, 120, 122, 124, 126 and 128. Bias current is applied to the diodes CRl and CR2 by means of a driver current I applied to terminal "a"
by means of inductances 130 and 132. In an identical fashion, bias current is applied to CR3 and CR4 by means of a driver current Ib applied to terminal "b" via the inductances 134 and 136. The circuit shown in Figure 5 is operable such that when CRl and CR2 are biased to have relatively low RF resistance, the diodes CR3 and CR4 are biased to have relatively high RF resistance and the bi-polar attenuator provides minimum attenuation at output terminal 138 without signal inversion. When diodes CRl and CR2, on the other hand, are biased to exhibit relatively high RF resistance, diodes CR3 and CR4 are biased to exhibit a relatively low RF resistance, and the bi-po]ar network provides minimum attenuation with signal inversion. ~hen all four diodes CRl, CR2, CR3 and CR4 are biased to have equa] RF resistance, the bridge is balanced and maximum attenuation is provided at terminal 138 which couples to either signal line 94 or 96 shown in Figure 4.
The terminal "a" and "b" of the bi-polar attenuators are connected to respective driver circuits 90 and 92, one of which is shown in Figure 6. As shown in Eigure 6, the driver circuit 90 includes an I driver control voltage input terminal 140 which receives a control signal from the I driver control output terminal 142 of the complex correlator shown in Figure 7. The driver ~L~81~7~
circuit 90 further inclucles a first pair of transistors Ql and Q2 coupled to a resistor-diode network 142 including diodes CR5, CR6, and CR7 and a second pair of transistors Q3 and Q4 coupled to a second resistor-diode network 144 including diodes CR8, CR9 and CR10. Transistors Ql and Q3 operate as emitter-follower transistors with the drive currents I and Ib emanating from the collectors of transistors Q2 and Q4, respectively. An operational amplifier 145 is inserted between the input terminal 140 and the emitter-follower Q3 in order to provide a unity again inverting amplifier and thus provide currents I and Ib whose variations are of mutually opposite polarity.
The resistor-diode networks 142 and 144 connected in the emitters of transistors Q2 and Q4 provide non-linear shaping of the control currents I
and Ib, respectively, to compensate for the non-linear control characteristics of the PIN diodes CRl ... CR4 of the bi-polar attenuator 86. The level of the I driver control input voltage applied to the input termina:L 140 controls the output currents of the drive transistors Q2 and Q~ in order to vary the RF
attenuation exhibited by the bi-polar attenuator 86. It should be noted, how-ever, that the resistor-diode networks 142 and 144 each include a variable resistor 146 and 148, respectively, which are used to set the current values of I and Ib for maximum RF attenuation when the amplitude of the con-trol voltage applied to terminal 140 is substantially mid-way in its input range, for example, if the input voltage varies from 1 to 5 volts, the resistors 146 and 148 are adjusted to provide for tnaximum attenuation when the control voltage is in the region of 2.5 to 3 volts. Additionally, transistors Q2 and Q4 are powered from a t 6V supply which is turned on only when the transmitter section 20 (Figure 2) is enabled. The circuit configurations for the bi-polar attenuator 88 and the driver 92 are identical and their operation is the same as described above and therefore need not be repeated.
Referring now to Figure 7, the I and Q driver control voltages for the complex weight circuit 66 shown in Figure 3 are derived by a weight control unit 68 which consists of a circuit which implements a complex correlation between the I and Q quadrature components of the reference signal REF taken ~From the coupler 102 and 104 (Figure 4), and the error signal taken from the coupler 72 shown in Figure 3. Accordingly, as shown in Figure 7, the in-phase I and quadrature Q components oE the reference signal REF from the complex weighting circuit are appliecl to input terminals 110 and 112 while the error signal from the coupler 72 is applied to input terminal 150. The error signal input is cormected to a power divider 152 which provides output signal lines 154 and 156 which respectively are connected to a pair of signal mixers 158 and 160 along with I and Q components of the reference signal REF. As shown, mixer 158 is adapted to correlate the I component while the mixer 160 is adapted to correlate the Q component. In addition, a DC bias is applied to the mixers 158 and 160 at termina]s 162 and 164, respectively, so that the out-put therefrom which appears on signal leads 166 and 168 rides on a DC :Level corresponding to the amplitude of the bias vo]tage applied to the mixers. The output of the I component mixer 158 is fed to an operational amplifier 170 which includes a low pass filter network 172 coupled in a feedback arrangement around the amplifier. Accordingly, the driver control signal for the in-phase I driver ciruit 90 shown in Figure 4 is coupled from output terminal. 142 to the I driver control input terminal 140 of Figure 4. Additionally, operational amplifier 170 includes a grounded variable resistance element 174 which pro~
vides a DC offset adjustment for the level of the output control voltage appearing at terminal 142. In a like manner, the quadrature Q component of the reference signal ~EF which is correlated in the mixer 160 is applied to an operational amplifier 176 having the filter feedback network 178 coupled thereto to provide a filtered amplified driver signal to the Q component driver 92 shown in Figure 4 by a suitable connection of terminal l44 and as shown in Figure 7 to terminal 141 shown in Figure 4. DC offset adjustment of the output from the amplifier 176 is provided by means of the grounded variable res:istance element 180. Also both operational amplifiers 170 and 176 are operated as differential amplifiers which have their respective + inputs connected to a reference voltage of ~3V applied t terminals 182 and 184, respectively, and are adapted to drive the output signal amplitude and phase . . .
7f~
from comple~ weighting ci.rcu:it 66 to the values needed to cancel the retrans-mitted signal.
Thus what has been shown and described is an i.n-band non-sampling real-time VHF radio repeater/relay tra[lscei.ver for use in two-way multi-hop remote sensor clata links whlch includes unique interference cancellation circuitry which operates to prevent the high level transmitted signal produced by the transmitter section of the transceiver from desensitizing the low noise RF amplifier located in the :Front of the receiver sectlon of the transceiver.
: While the invention has been particularly shown and described with reference to a preferred embodiment thereof, it is to be understood from the foregoing that other changes and modifications in the form and details may be made without departing from the spirit and scope of the invention as set forth in the subtended claims.
Claims (16)
PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:
1. Radio apparatus for operating as a signal repeater for the relaying of information signals from one location to another over a network, comprising:
co-located receiver and transmitter means coupled to a common antenna and being operable such that RF signals received by said receiver means are retransmitted without substantial delay from said transmitter means and wherein said receiver means includes RF input amplifier means, and further includes a first adaptive interference signal cancellation means coupled between said common antenna and said RF input amplifier means for preventing RF signals being currently retransmitted from rendering said receiver means unresponsive to concurrently received RF
signals, and further including a second adaptive interference signal cancellation means coupled to the output of said RF input amplifier means to cancel any residue of interferrence signals not cancelled by said first adaptive interference signal cancellation means, whereby said receiver and transmitter means may be simultaneously operated without signal interference between one another thereby permitting near continuous use of both said receiver and transmitter means at a substantially 100% duty cycle.
co-located receiver and transmitter means coupled to a common antenna and being operable such that RF signals received by said receiver means are retransmitted without substantial delay from said transmitter means and wherein said receiver means includes RF input amplifier means, and further includes a first adaptive interference signal cancellation means coupled between said common antenna and said RF input amplifier means for preventing RF signals being currently retransmitted from rendering said receiver means unresponsive to concurrently received RF
signals, and further including a second adaptive interference signal cancellation means coupled to the output of said RF input amplifier means to cancel any residue of interferrence signals not cancelled by said first adaptive interference signal cancellation means, whereby said receiver and transmitter means may be simultaneously operated without signal interference between one another thereby permitting near continuous use of both said receiver and transmitter means at a substantially 100% duty cycle.
2. The radio apparatus as defined by Claim 1 wherein said receiver means and said transmitter means have respective operational frequencies which are offset with respect to one another.
3. The radio apparatus as defined by Claim 2 wherein said receiver means includes circuit means for down-converting said received RF signals of a predetermined frequency band to IF
signals, and wherein said transmitter means includes circuit means for being coupled to said IF signals and including means for up-converting said IF signals to RF signals having a frequency which is in the same said frequency band but offset with respect to the frequency of said received RF signals.
signals, and wherein said transmitter means includes circuit means for being coupled to said IF signals and including means for up-converting said IF signals to RF signals having a frequency which is in the same said frequency band but offset with respect to the frequency of said received RF signals.
4. The radio apparatus as defined by claim 3 wherein said transmitter means includes variable attenuator means coupled to said antenna for selectively varying the power of the retransmitted RF signals from said antenna.
5. The radio apparatus as defined by claim 3 wherein said receiver means additionally includes received RF signal presence detector means which is operable to operationally enable said transmitter means upon receiving RF signals at said receiver means.
6. The radio apparatus as defined by claim 1 wherein said transmitter means retransmits received RF signals at a different frequency in the same frequency band from the frequency of the received RF signals.
7. The radio apparatus as defined by claim 6 wherein the frequency of the received RF signals and the frequency of the retransmitted RF signals are in the VHF frequency band.
8. The radio apparatus as defined by claim 1 wherein said first adaptive interference signal cancellation means includes:
first circuit means for providing a composite input RF
signal including the received RF signal from said antenna along with any undesired cross-coupled RF signals being retransmitted by said transmitter means and any RF signals reflected from the surrounding locality to said antenna;
second circuit means for providing an RF reference signal from the RF signals being coupled to said antenna for retransmission by said transmitter means;
third circuit means for adaptively adjusting the phase and amplitude of said RF reference signal to effect cancellation of said undesired cross-coupled RF signals when combined with said composite input RF signal; and fourth circuit means coupled to said first and third circuit means for algebraically combining said composite input RF
signal and the adaptively adjusted RF reference signal to substantially effect said cancellation.
first circuit means for providing a composite input RF
signal including the received RF signal from said antenna along with any undesired cross-coupled RF signals being retransmitted by said transmitter means and any RF signals reflected from the surrounding locality to said antenna;
second circuit means for providing an RF reference signal from the RF signals being coupled to said antenna for retransmission by said transmitter means;
third circuit means for adaptively adjusting the phase and amplitude of said RF reference signal to effect cancellation of said undesired cross-coupled RF signals when combined with said composite input RF signal; and fourth circuit means coupled to said first and third circuit means for algebraically combining said composite input RF
signal and the adaptively adjusted RF reference signal to substantially effect said cancellation.
9. The radio apparatus as defined by claim 8 wherein said third circuit means includes means for adjusting the phase and amplitude of said RF reference signal in response to an error signal, and wherein said fourth circuit means includes means for generating said error signal in response to the summation of said composite input RF signal and said adaptively adjusted RF
reference signal, and means for feeding said error signal back to said third circuit means.
reference signal, and means for feeding said error signal back to said third circuit means.
10. The radio apparatus as defined by claim 9 wherein said third circuit means includes: complex weighting circuit means and complex weighting control circuit means for controlling said weighting circuit, said control circuit means being responsive to said error signal and said RF reference signal provided by said second circuit means to provide control signals to said weighting circuit.
11. The radio apparatus as defined by claim 10 wherein said complex weighting circuit includes means coupled to said second circuit means for developing in-phase and quadrature component signals from said RF reference signal, first and second bi-polar attenuator circuit means and respective driver circuit means therefore coupled to said in-phase and quadrature component signals, said driver circuit means being coupled to and responsive to control signals from said complex control circuit means to vary the attenuator characteristic of said first and second bi-polar attenuator circuit means in response to said error signal, and combiner circuit means coupled to the outputs of said first and second bi-polar attenuator circuit means and circuit means coupling said combiner circuit means to said fourth circuit means whereby cancellation is effected by summing the output signal of said combiner circuit means with said composite input RF signal.
12. The radio apparatus as defined by claim 11 wherein said complex weighting control circuit means comprises a complex correlator including means responsive to said error signal to provide in-phase quadrature component error signals therefrom, first and second signal mixers being respectively coupled to said in-phase and said quadrature component error signals together with said in-phase and said quadrature component signals of said RF
reference signal to provide an in-phase component control signal and a quadrature component control signal for respectively controlling said first and second bi-polar attenuator circuit means.
reference signal to provide an in-phase component control signal and a quadrature component control signal for respectively controlling said first and second bi-polar attenuator circuit means.
13. The radio apparatus as defined by claim 12 and additionally including first and second amplifier means respectively coupled to the output of said first and second signal mixers for providing in-phase and quadrature control signals respectively to driver circuit means coupled to said first and second bi-polar attenuator circuit means.
14. The radio apparatus as defined by claim 13 and addditionally including lowpass filter circuit means coupled to said first and second amplifier means.
15. The radio apparatus as defined by claim 14 wherein said transmitter means is operative at a different RF frequency in the same frequency band from the operative RF frequency of said receiver means.
16. The radio apparatus as defined by claim 15 wherein said frequency band comprises the VHF frequency band.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US06/325,162 US4406016A (en) | 1981-11-27 | 1981-11-27 | VHF Sensor in-band radio relay |
| US325,162 | 1981-11-27 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1188740A true CA1188740A (en) | 1985-06-11 |
Family
ID=23266697
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000416626A Expired CA1188740A (en) | 1981-11-27 | 1982-11-23 | Vhf sensor in-band radio relay |
Country Status (2)
| Country | Link |
|---|---|
| US (1) | US4406016A (en) |
| CA (1) | CA1188740A (en) |
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| CA1250022A (en) * | 1985-09-18 | 1989-02-14 | Nec Corporation | One frequency repeater having interference cancellation capability in a digital radio system |
| CA1333413C (en) * | 1988-07-13 | 1994-12-06 | Andre Gagnon | Frequency independent strong signal suppressor |
| US10361802B1 (en) | 1999-02-01 | 2019-07-23 | Blanding Hovenweep, Llc | Adaptive pattern recognition based control system and method |
| US8352400B2 (en) | 1991-12-23 | 2013-01-08 | Hoffberg Steven M | Adaptive pattern recognition based controller apparatus and method and human-factored interface therefore |
| US5689267A (en) * | 1995-11-22 | 1997-11-18 | Allied Signal Inc | Automatic VSWR sensing for aircraft-mounted pulse radar systems |
| US6049720A (en) * | 1996-04-12 | 2000-04-11 | Transcrypt International / E.F. Johnson Company | Link delay calculation and compensation system |
| US5896560A (en) * | 1996-04-12 | 1999-04-20 | Transcrypt International/E. F. Johnson Company | Transmit control system using in-band tone signalling |
| US5991309A (en) * | 1996-04-12 | 1999-11-23 | E.F. Johnson Company | Bandwidth management system for a remote repeater network |
| AU2662997A (en) * | 1996-04-12 | 1997-11-07 | E.F. Johnson Company | Transmit control system using control tone signalling |
| US7054271B2 (en) | 1996-12-06 | 2006-05-30 | Ipco, Llc | Wireless network system and method for providing same |
| US8982856B2 (en) | 1996-12-06 | 2015-03-17 | Ipco, Llc | Systems and methods for facilitating wireless network communication, satellite-based wireless network systems, and aircraft-based wireless network systems, and related methods |
| US6665009B1 (en) * | 1998-05-20 | 2003-12-16 | Omnivision Technologies, Inc. | On-chip dead pixel correction in a CMOS imaging sensor |
| US6891838B1 (en) | 1998-06-22 | 2005-05-10 | Statsignal Ipc, Llc | System and method for monitoring and controlling residential devices |
| US6914893B2 (en) | 1998-06-22 | 2005-07-05 | Statsignal Ipc, Llc | System and method for monitoring and controlling remote devices |
| US8410931B2 (en) | 1998-06-22 | 2013-04-02 | Sipco, Llc | Mobile inventory unit monitoring systems and methods |
| US6437692B1 (en) | 1998-06-22 | 2002-08-20 | Statsignal Systems, Inc. | System and method for monitoring and controlling remote devices |
| US8364136B2 (en) | 1999-02-01 | 2013-01-29 | Steven M Hoffberg | Mobile system, a method of operating mobile system and a non-transitory computer readable medium for a programmable control of a mobile system |
| US7966078B2 (en) | 1999-02-01 | 2011-06-21 | Steven Hoffberg | Network media appliance system and method |
| US7650425B2 (en) | 1999-03-18 | 2010-01-19 | Sipco, Llc | System and method for controlling communication between a host computer and communication devices associated with remote devices in an automated monitoring system |
| US6826607B1 (en) | 1999-10-06 | 2004-11-30 | Sensoria Corporation | Apparatus for internetworked hybrid wireless integrated network sensors (WINS) |
| US7020701B1 (en) | 1999-10-06 | 2006-03-28 | Sensoria Corporation | Method for collecting and processing data using internetworked wireless integrated network sensors (WINS) |
| US6859831B1 (en) | 1999-10-06 | 2005-02-22 | Sensoria Corporation | Method and apparatus for internetworked wireless integrated network sensor (WINS) nodes |
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| US7484008B1 (en) | 1999-10-06 | 2009-01-27 | Borgia/Cummins, Llc | Apparatus for vehicle internetworks |
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| US6832251B1 (en) | 1999-10-06 | 2004-12-14 | Sensoria Corporation | Method and apparatus for distributed signal processing among internetworked wireless integrated network sensors (WINS) |
| ES2160087B1 (en) * | 2000-02-18 | 2003-03-01 | Mier Comunicaciones S A | PROCEDURE FOR REPETITION OF SIGNALS IN INSOFREQUENCY AND REPEATER OF SIGNS IN ISOFREQUENCY. |
| US7480501B2 (en) | 2001-10-24 | 2009-01-20 | Statsignal Ipc, Llc | System and method for transmitting an emergency message over an integrated wireless network |
| US8489063B2 (en) | 2001-10-24 | 2013-07-16 | Sipco, Llc | Systems and methods for providing emergency messages to a mobile device |
| US7424527B2 (en) | 2001-10-30 | 2008-09-09 | Sipco, Llc | System and method for transmitting pollution information over an integrated wireless network |
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| US8031650B2 (en) | 2004-03-03 | 2011-10-04 | Sipco, Llc | System and method for monitoring remote devices with a dual-mode wireless communication protocol |
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| CN105119629B (en) * | 2015-07-10 | 2018-04-06 | 惠州Tcl移动通信有限公司 | Co-channel full duplex terminal and system simultaneously |
| CN108023608B (en) * | 2017-12-04 | 2020-03-06 | 中兴通讯股份有限公司 | Full time duplex system, circuit, terminal, control method and readable medium |
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|---|---|---|---|---|
| US3566234A (en) * | 1968-11-07 | 1971-02-23 | Cutler Hammer Inc | Concurrent same-frequency fm radio repeater |
| US3611139A (en) * | 1970-02-25 | 1971-10-05 | Us Navy | Orthogonal mixer f{11 {0 f{11 {0 repeater |
| US3696429A (en) * | 1971-05-24 | 1972-10-03 | Cutler Hammer Inc | Signal cancellation system |
| JPS55133148A (en) * | 1979-04-04 | 1980-10-16 | Nippon Telegr & Teleph Corp <Ntt> | Interference wave removal system |
| US4317217A (en) * | 1980-08-11 | 1982-02-23 | Motorola, Inc. | Tag generator for a same-frequency repeater |
-
1981
- 1981-11-27 US US06/325,162 patent/US4406016A/en not_active Expired - Fee Related
-
1982
- 1982-11-23 CA CA000416626A patent/CA1188740A/en not_active Expired
Also Published As
| Publication number | Publication date |
|---|---|
| US4406016A (en) | 1983-09-20 |
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|---|---|---|---|
| MKEC | Expiry (correction) | ||
| MKEX | Expiry |